Figure 1: Inexpensive PT2399-based audio delay board as found on the usual Internet sites. Click on the image for a larger version.

In an earlier blog post (Fixing the CAT Systems DL-1000 and PT-1000 repeater audio delay boards - LINK) I discussed the modification of a PT2399-based audio delay line for use with the CAT-1000 repeater controller - and I also hinted that it would be possible to take an inexpensive, off-the-shelf PT2399-based audio effects (echo/reverb) board and convert it into just a delay board. 

While the uses of an echo-less delay for more mundane purposes may be apparent, it would be fair to ask why might one use an audio delay in an amateur radio repeater?  There are several possibilities:

• The muting of DTMF ("Touch Tone") control signals.  Typically, it takes a few 10s of milliseconds to detect such signals and being able to delay the audio means that they can be muted "after" they are detected.
  

• Reducing the probability of cutting off the beginning of incoming transmissions due to the slow response of a subaudible tone.  By passing COS-squelched audio through the delay - but gating it after the delay, one may still get the benefits of a tone squelch, but prevent the loss of the beginning of a transmission.  This is particularly important on cascaded, linked systems where it may take some time for the system to key up from end-to-end.
  

• The suppression of squelch noise burst at the end of the transmission.  By knowing "before-hand" when an input signal goes away, one can mute the delayed audio such that the noise burst is eliminated.
  

Making good on the threat in the previous article, I reverse-engineered one of the PT2399-based boards available from Amazon and EvilBay and here, I present this modification, using one of these boards as a general-purpose audio delay.

The board:

Figure 2: Schematic diagram of the audio delay board, with modification instructions. This diagram is reverse-engineered from the board depicted in Figure 1. Click on the image for a larger version.

The PT2399 boards found at the usual Internet sellers like EvilBay or Amazon are typically built exactly from the manufacturer's data sheet, and one of those found on the Internet for less than US$10 is depicted in Figure 1.  (Note that the chip may have another prefix in front of the number, such as "AD2399" or "CD2399")

The pictured board is surprisingly well-built, with plenty of bypassing of the voltage supply rails and a reasonable layout.  Despite the use of small, surface-mount resistors, it is fairly easy to modify, given a bit of care, and most of the components have visible silkscreen markings, making it easy to correlate the reverse-engineered circuit diagram (above) with the on-board components.

A few of the components do not have visible silkscreen markings (perhaps located under the components themselves?) and these are labeled in the circuit diagram and the board layout diagram (below in Figure 3) with letters such as "CA", "CB", "RA", etc.

Figure 3:  Board layout showing component designations of the board in Figure 1. Note that some of the components have no silkscreen markings and are labeled with letters that have been arbitrarily identified as "CA", "CB", "RA", etc. Click on the image for a larger version.

Removing the echo, making it delay-only

This circuit is the "bog standard" echo/reverb circuit from the app note - but it requires modification to be used as a simple audio delay as follows:

• The output audio needs to be pulled from a different location (pin 14 rather than pin 15):
• Remove R22, the 5.6k resistor in series with the output capacitor marked "CC".
• A jumper needs to be placed between the junction of the (former) R22 and capacitor "CC" and pin 14 of the IC as depicted in Figure 4, below.
  

• The feedback for the reverb need to be disabled and this involves the removal of capacitors C15 and C17.

Figure 4: The modified PT2399 board, showing the jumper on pin 14 and the two flying resistors on the potentiometer, now used for delay adjustment.  Note the deleted C15 and C17. Click on the image for a larger version.

Figure 5, below, shows the schematic of the modified board with the changes described above.

At this point the board is converted to being a delay-only board, but with the amount of delay fixed at approximately 200 milliseconds with the value of R27  being 15k as seen in table 1, below.  This amount of delay is quite reasonable for use on a repeater to provide the aforementioned functions with no further modifications.

Optional delay adjustment:

By removing the need to be able to adjust the amount of echo/reverb, we have freed the 50k potentiometer, "RA", to be used as a delay adjustment as follows:

• Remove R27, the 15k resistor, and replace this with a 47k resistor.  This is most easily done by using a 1/4 or 1/8 watt through-hole resistor and soldering one end directly to pin 6 and the other to ground, using the middle "G" pin along the edge of the board.
  
• Remove R21 and using a 1/4 or 1/8 watt leaded 4.7k resistor, solder one end across where R21 went (to connect the wiper of potentiometer "RA") to pin 6 of the IC.
• The 4.7k resistor (and parallel 47k resistor) sets the minimum resistance at about 4.3k while the maximum resistance is set by the parallel 47k resistor and the 50k potentiometer in series with the 4.7k resistor at about 25.3k.  These set the minimum and maximum delay attainable by adjustment of the potentiometer.
  

Of course, one may also use surface-mount resistors rather than through-hole components, using jumper wires rather than the flying leads of the components. 

Figure 5:  Diagram of of the '2399 board after the modifications to be a "delay-only" circuit. Click on the image for a larger version

This modification provides a delay that is adjustable from a bit more than 300 milliseconds to around 80 milliseconds, adjustable via the variable potentiometer. 

It's worth noting that if you do NOT  require a variable delay, using fixed resistors may offer better reliability than an inexpensive potentiometer of unknown quality - something to consider if the board is to be located on a remote repeater site.

If variable delay is not required, one would not use the 4.7k resistor at the junction of R21/"RA" - or use the potentiometer at all, and R27 would be replaced with a fixed resistor, the value chosen for the desired amount of delay as indicated in the following table:

The chart above shows examples of resistance to attain certain delays, but standard resistor values may be used and the amount of delay interpolated between it and the values shown in the table.  

While not specified in the data sheet, the delay will vary with temperature to a slight degree as the onboard oscillator drifts, so it is recommended that the needed delay be chosen such that it will allow a slight variance while still providing the amount of delay for the needed task.

Comment: 

If this is to be powered from a 12 volt supply, it's suggested that one place a resistor in series with the "+" input to provide additional decoupling of the power supply.  The (possible) issue is that the 470uF input capacitor ("CA" on the diagram) will couple power supply noise/ripple into the ground of the audio delay board itself - and associated audio leads - potentially resulting in circulating currents (ground loop) which can induce noise.  Additionally, an added series resistance provides a modicum of additional protection against power supply related spikes.

The board itself draws less than 50 milliamps, and as long as at least 8 volts is present on the input of U4, the 5 volt regulator, everything will be fine.  A 1/4-watt 47 ohm resistor (any value from 33 to 62 ohms will work) will do nicely. 

* * * * * * *

Addendum:  Adding audio switching

Since the original publication of this post there have been several questions as to how to "switch" audio to the delay board.  In many cases, this will not be required as the device being used (say, a repeater controller) may already have an audio gate - but in the event that you really do need to switch audio on/off - or switch it between "A" and "B", refer to Figure 6, below.

Figure 6: Examples of using the 4066 quad audio gate for audio gating and switching. Both an "on/off" gate and "A/B" switch - plus using a 4066 to generate an inverted logic signals - is depicted. Click on the image for a larger version.

How it works:

For the audio switching we will use the 4066 quad analog switch.  In this example, we are using the CD4066 - the "old school" 4000-series CMOS which can operate between 3 and 15 volts.  The "newer" "HC" logic versions may also be used, but their maximum voltage is either 5 or 6 volts, depending on the specific part used. 

The "On/Off" gate:

Let's take the On/Off gate as the first example.  Note that the input/output ports - which are interchangeable (e.g. the switch is bidirectional so it could even be used with bidirectional signals) - are biased with R201 and R202 which sets the resting DC voltage at about 1/2 the supply voltage from the circuit marked "V+/2 Source".  Capacitors are used on these lines to block this DC bias voltage from appearing on the In/Out lines and disrupting the bias.  If you are switching audio lines with DC already on them, be sure to consider the polarity of the blocking capacitors in the event that this "external" audio source's voltage is higher than V+/2.

The reason for adding a bias voltage to the In/Out audio is to prevent the audio swing from causing the protection diodes found on this (and almost all other) chips from conducting if it exceeds either V+ or goes "below" ground:  Doing so would likely cause distortion of the audio on the positive and/or negative peaks.

Note that the bias is applied to both the input and output.  This is done to prevent an audio "click" or "pop" that would occur when the switch was closed:  If the DC voltages weren't exactly equal on the in/out lines when the switch was open, closing (turning on) the switch would cause a sudden change in the form of a click.

The "A/B" gate:

If you wish to switch two different audio signals from the same logic signal by turning one or the other on, this circuit is a replication of the "On/Off" gate - but it uses another 4066 gate as a logic inverter.  When the "A" switch is on, U1d - the middle switch - is also turned on, shorting R303 to ground which turns of the "B" switch.  When the "A" switch is turned off by setting its logic level to low, U1d is now turned off but the control line for the "B" switch is pulled high by R303, turning it on.

While the example shows two separate switches, one could connect them together, tying one of the in/out lines of each switch together as the common in/out port if you wished to use it to select source "A" or source "B".  If you do this, you could probably eliminate one of the blocking capacitors - but there's little harm if leaving it there if you are unsure as to what to do.

The "Low Voltage Logic to High Voltage Logic" converter:

All digital ICs have threshold voltages for their logic inputs - and the 4066 is no exception.  If you operate the 4066 gates from 12 volts, you will need "about" 12 volts on the "control" pin to properly "turn on" the audio gate:  Applying, say, 5 volts to it as a "high" signal probably won't work so the voltage of this control signal must match the supply voltage of the switch chip.

This is a very simple one-transistor logic level converter.  In the event that you have, say, a repeater controller that has 3.3 volt logic, but you choose to power the 4066 audio switches from 12 volts, you can use this to derive the 12 volt logic level needed to properly switch.  One downside of this circuit is that it will "invert" the logic signal:  Input a "1" (high voltage) and you get a "0" (low voltage) on the output.

Depending on the audio control signal from your controller, it may already be a "low active" type - or it may be programmable.  In the event that you need to do a high voltage logic level and  that it NOT be inverted you can put two of these one-transistor circuits in series.  If you are already needing to switch between audio "A" and "B", you wouldn't need to do this as you could simply swap "A" and "B" if you end up with an "inverted" control signal.

Selection of power supply voltage:

As mentioned, the CD4066 may operate from anywhere from 3 to 15 volts:  12 volts is sometimes convenient as that may be the unregulated input voltage of the main power supply - but what voltage is appropriate?

The supply voltage should be equal to or higher than the peak-to-peak audio signal - something that can only be measured accurately with an oscilloscope.  For example, if you have a repeater and the peak audio voltage from the audio line when the receiver is running open squelch with no signal is 8 volts, you should NOT power the 4066 audio gate from 5 volts - but 10 or more volts would certainly provide adequate headroom.  If your audio level peak-to-peak voltage exceeds the power supply voltage, the audio will be clipped by the 4066's protection diodes and cause audio distortion.

If, in the above example, the peak voltage from the squelch noise was only 3.5 volts peak-to-peak, you could operate the 4066 from a 5 volt supply, saving you the need for logic level conversion and alsopermitting the use of the "74HC4066" instead.

Consideration of impedance:

These switches are intended for "high" load impedance (typically 10k or more) audio input rather than for audio switching where the LOAD impedance is low - such as a speaker.  The reason for this has to do with the resistance of the 4066 gates (which could be 10s or 100s of ohms) and, to a lesser extent, the value of the blocking capacitors  Fortunately, the input impedance of most sources on which this would be used (audio amplifier, repeater controller) are typically quite high.

* * * * * * *

This page stolen from ka7oei.blogspot.com

[END]

In the mid 90s I decided to throw together what I called a "Bat Listener" - a simple receiver used to convert ultrasonic sound down to the audible range.

Figure 1: The exterior of the ultrasonic receiver, complete with fancy labeling! Click on the image for a larger version.

Two types of circuits:

There are two common ways to convert a higher (ultrasonic) signal to the audible range, whether this is done using analog or DSP (Digital Signal Processing) techniques.

Frequency division

There are several ways to do this, the simplest being the "divider" type which digitally converts ultrasonic frequencies to audible by integer division of the input to a lower frequency.

The problem with this simple approach is that it does not preserve the amplitude (loudness) of the original sound since it must take the input signal, amplify/convert it to a series of logic-level pulses - which loses any amplitude reference - and do a brute-force digital division.  Additionally, if there are multiple signals present, for the most part only the strongest one will be converted down.

Clearly, one cannot "tune" this type of circuit:  A signal at 40 kHz will always be divided down by a fixed integer amount,  Let's say that the circuit digitally divides by 32:  That 40 kHz signal will be at 1.25 kHz.

Additionally, the direct "A-B" frequency differences between ultrasonic signals is lost, instead being "(A-B)/N" where "N" is the number of divisions.  In other words, the relative frequency differences between signals is not preserved.

Heterodyne conversion

The other way to do this is to convert the frequency.  In this technique, two signals - the ultrasonic to be converted - and another generated by the device (the "local" oscillator) are mixed together.  The result is an arithmetic shift in frequency.

The biggest advantages of this method are the fact that that not only are the differences in frequency preserved (e.g. two tones 1 kHz apart at ultrasonic will appear as two tones 1 kHz apart at audio) but the relative amplitudes (loudnesses) of the received signals are preserved as well.

Frequency conversion:

I chose to build a heterodyning receiver to convert the input frequency to a lower one.  This can preserve the amplitude and frequency relationships  - plus it is fully tunable, allowing one to choose the frequency range to convert to audible sounds - and since it is a simple conversion, multiple signals present will also be preserved.

When it comes to frequency conversion, there are two ways:  The simplest - direct conversion - would involve mixing a variable oscillator with the incoming signal and filtering/amplifying the resulting audio.  This has the advantage of being the easiest, and it is the method described in this article:

     April, 2006 QST article, A Home-made Ultrasonic Power Line Arc Detector - link)

While I could have easily built something like this a decade before the above article was published, as I'm sometimes wont to do I decided to make it a bit more complicated, constructing a superheterodyne converter.

While a direct-conversion receive simply mixes an oscillator with the desired signal to cause a frequency conversion, a superheterodyne receiver operates like a conventional AM or FM radio:  The desired signal is first converted to an IF (Intermediate Frequency) - and this IF is then converted to audio.  The advantage of the superheterodyne scheme is that filtering may be applied at the IF to limit the receive bandwidth - and since the IF is fixed, its width remains constant over the tuning range, just like that in a conventional radio/receiver.

Circuit description

Figure 2: Schematic diagram of the superheterodyne ultrasonic receiver. See text for a circuit description. Click on image for a larger version.

As noted above, this circuit is more complicated than it needs to be, so make of it what you will!

VCO:

The heart of the unit is U1, the VCO (Voltage Controlled Oscillator) which uses the venerable CD4046 PLL chip.  Often used for frequency synthesis, we are using (only) the oscillator portion, which provides a linearly-tuned and fairly stable frequency source, adjusted by the voltage applied via R101 (and scaling resistor R102).  The values were chosen to provide an approximate frequency range of 125 to 185 kHz (more on this later) to allow tuning of audio signals from (ostensibly) 0 to about 60 kHz.  The actual tuning range is closer to 115-190 kHz as a bit of extra margin for the frequency range.

The only critical component here is C101 which should be a frequency-stable capacitor.  I used a polystyrene capacitor, but an NP0 (a.k.a. C0G) or silver-mica could be used, instead.  When I reverse-engineered this device, I noted that the marked capacitance value was unreadable, but back-of-the-envelope calculates indicate that a value of "about 150pf" should be in the ballpark.

R103, connected to the "R1" pin of U1, sets the approximate center frequency range while R104, connected to the "R2" pin - sets the lowest frequency - which important, since we want to constrain the tuning to 125-185 kHz.  Additionally, the low end of the tuning range was further refined by R102 on the "ground" side of the tuning potentiometer, which sets the minimum voltage that may be applied to the "VCOIN" pin.

The VCO output, a square wave, is buffered by U2, a hex inverter, and several sections are used to provide both a VCO signal and its inverted version to drive the mixer.

While the 4000 series CMOS chips throughout this receiver will happily run from 3-15 volts, they are operated from a regulated 5 volt supply - mainly to improve frequency stability and to provide a nice, stable voltage for a few other low-level circuits and to provide isolation from the main battery supply which will vary a bit, particularly at higher receive volumes:  This variance, if it gets back into some earlier stages, could cause instability of the receiver in the form of "motorboating" or some other type of feedback.

BFO:

Another circuit is the BFO (Beat Frequency Oscillator) which is used to convert the IF signal back down to audio - both being processes that we'll discuss shortly.  This uses an inexpensive 500 kHz ceramic resonator to form an oscillator using one of the sections of U2C, the signal being buffered by U2B.  This signal is divided-by-two using U3A, one half of a 4013 dual flip-flop - and then divided by two again using U3B, yielding a stable 125 kHz signal.  As with the VCO, two phases of this signal (normal and inverse) are available, this time using the "Q" and "!Q" outputs of the 4013.

Input signal path:

J1, a disconnect-type 3.5mm stereo jack is wired so that an internally-mounted electret "capsule" microphone is connected by default.  This microphone element (M301) is of the "2 wire" type or electret microphone in which a bias voltage is applied to the same pin from which audio is drawn - this voltage being applied via R301 from the 5 volt regulated supply.  The specific make/model of this electret element is unknown as it was selected from a small collection to find the best performer at ultrasonic.

At some point in the future, I'll replace this with a more modern MEMs microphone as described in Another article:  Improving my ultrasonic sniffer for finding power line arcing by using MEMs microphones - link.

The signal from the microphone is applied to U4A which is wired as a unity-gain buffer.  For this, an LM833 is used, an inexpensive, low-noise dual op amp:  An LM358 or many other types may be used here as well - just make sure that it is is fairly low noise:  I'd avoid the use of the LM1458 here as it is quite noisy by comparison!

Section U4B amplifies the signal voltage by 10 (20 dB of power gain) and this signal is applied via R305 to a simple L/C high-pass filter consisting of C303, C304, L301 and L302 the latter two components being inexpensive 18 milliHenry inductors.  Certainly, an R/C-based high-pass filter could have been constructed using U4B, but I chose not to do that for some reason.

Figure 3: Inside the ultrasonic receiver, constructed on prototype board and having been modified several times over the years. Click on the image for a larger version.

In simulation, the C303/C304/L301/L302 filter has a -3dB roll-off of about  23 kHz, it's down by 10dB at about 19.5 kHz, by 20dB at about 16 kHz and by 40 dB at 9 kHz and with the values shown, it's flat to within 1 dB between about 24 and 100 kHz.

The output of the filter is amplified by U5B - and then even more by U5A (which has a bit of roll-off from C307) to yield a whole lot of gain.  It's very possible that I over-did the gain here, but unless the signal source is quite close, there is no clipping observed on the output of U5A.

Its worth noting that a mid-supply voltage is created using R309/R310 to provide a "virtual ground" for the op amps and to maintain stability, it is heavily filtered by C306 and C302, each located near the respective op amp shown on the diagram.

Mixer and band-pass filter:

It is this next section that may seem unfamiliar to some - the use of a CMOS analog switch as a signal mixer.  For this, a CD4066 is used which consists of four separate analog switches.  The filtered and amplified ultrasonic input signal from U5A via C308 is applied to pins 2 and 10 of U6A/U6D.  When the respective signals on the control pins "VCO_A" and "VCO_B" go high, the switches are activated, and because VCO_A and VCO_B are inverts of each other, each of these switches is closed in turn.  The result of this is that the inputted signal is chopped up at the rate of the 125-185 kHz VCO and this produces two mixing products.  

For example, let's assume that there is a 40 kHz signal is present on the input that we wish to hear.  If the VCO is tuned 40 kHz above the 125 kHz IF (again, more on that momentarily) - to a frequency of 165 kHz - the switching action of U6A and U6D produces both the sum (165 + 40 = 205 kHz) and the difference (165 - 40 = 125 kHz).

T301 is a filter/transformer that passes only the 125 kHz signal - the difference signal in this case.  This transformer consists of two separate windings, each resonated using its internal capacitors and the externally-added 820 pF capacitors on each winding (e.g. C309/C310) to "pad" it down to 125 kHz.  This forms a fairly wide (8-10 kHz) filter that rejects signals outside the immediate vicinity of its 125 kHz frequency.  Because this filtering is at a fixed frequency, it does not vary with input tuning which means that its bandwidth is constant over frequency.

Of all of the components in this device, this transformer is unique:  It was originally a 262.5 kHz IF transformer from a 1970s/1980s Philco (Ford) AM-only car radio.  While I could have certainly used the original 262.5 kHz frequency - or even 250 kHz, when I built this I decided to pad it down to 125 kHz using C309/C310  - a frequency that is conveniently 1/4th of the 500 kHz resonator.

It's been so long since I built this, I don't recall why I didn't simply divide the 500 kHz by two and readjust that transformer to 250 kHz.  Practically speaking, I could have also up-converted to 455 kHz and used either transformers or ceramic filters from a modern AM radio as 455 kHz ceramic resonators were certainly available at the time - but I didn't do that.

Each half of T301 has a center tap and to this, a bias voltage is applied via R315 to assure that the voltage on these switches was in the middle of the supply range, away from the protection diodes on the 4066's I/O pins, which could cause clipping/distortion should they be allowed to conduct if the signal voltage got too near the ground or supply rails.  To prevent coupling between the two halves of the transformer via the center tap, R314/C311 was added, the resistor adding isolation with the capacitor bypassing the remainder of the signal.  Practically speaking, being able to adjust the bias voltage was unnecessary as a simple resistive voltage divider to set the bias at 2.5 volts (1/2 the supply voltage) would have been just fine.

On the "other" side of the transformer is the other half of U6 (e.g. U6B/U6C) - this time, clocked from the fixed 125 kHz oscillator.  From this, the signal - previously converted up to 125 kHz is now converted back down to audio.

Post-mixer amp/LPF:

The output of the down-converting mixer is applied to U7B via R316, a 1k resistor and a 0.001uF capacitor, both of which form a simple R/C low-pass filter to attenuate any high-frequency leakage signals from the mixer.  Because the mixing process itself is a bit lossy (about 25% efficient) as is transformer/filter T301, U7B boosts the signal by a factor of 10 (20dB) and then applies it to U7A, which is configured as a variable gain amplifier section.  The output of this is then boosted again by U8, an LM386 which is capable of driving headphones or even a small speaker.

A few comments about the design:

Originally, the circuit lacked U7 at all, but it was added when the gain of U8 (the audio amplifier), by itself, was found to be inadequate.  Since U7 was "patched" into place, this explains the odd gain distribution:  If I were rebuilding this from scratch, I'd certainly not need two post-mixer amplifier sections and I could have likely eliminated one full dual op-amp package.  As it is, I may add a "high/low" gain switch somewhere around U5 to allow reduction of the gain somewhat when in the presence of possibly-high ultrasonic signal levels to prevent clipping prior to the band-pass filter which would surely degrade overall performance.

If I were to build this again I would likely use a 455 kHz IF, instead.  While not as plentiful, 455 kHz ceramic resonators are available to use for the BFO as are either transformer or ceramic-based band-pass filters.  I would also likely reconfigure U4B or U5 to perform the high-pass filter function rather than using harder-to-find inductors.

Again, I built this unit in the mid 1990s and have since lost my original notes, but I do recall that I modified it a few times since, simply tacking changes onto the old circuit rather than completely revising it.

Use as a longwave receiver:

While primarily intended to "hear" ultrasonic sounds such as those produced by bats, insects, leaking pipes, arcing power lines, etc., it is just a longwave radio receiver connected to a microphone:  If one connects a few 10s of feet/meters of wire to to J1 - and provides an Earth/ground reference to its shield connection - one can easily tune in the high-power transmitters used for submarine communications (around 20-30 kHz) plus the WWVB time signal at 60 kHz.  This must, of course, be done away from man-made noise sources such as power lines.

Alternatively, I have used a loop of about 1 foot (25cm) diameter of a dozen or so turns of wire along with a 10uF capacitor in series (to optionally block DC from R301) and been able to hear such signals - even in suburbia - but with this arrangement you'll also likely hear plenty of similar signals from the myriad switching supplies that likely inhabit your house as well!

Final comments:

The reader should be under no illusion that this is an optimized circuit or that I would do it this way again:  It was assembled fairly quickly to suit a need and to test a few random ideas, just to see if they would work.  Will I rebuild it at some point?  I don't know - it works as it should, so I don't plan to re-make something that is currently fit for purpose.

While I've heard very few bats with this - probably due to the deficiencies of the electret microphone at ultrasonic frequencies (which explains the future switch to MEMS-type microphones) - I've used it to find powerline noise (arcs are noisy at ultrasonic) and to test longwave receive antennas.

This page stolen from ka7oei.blogspot.com

[End]

At the Northern Utah WebSDR (link) we run a number of KiwiSDR receivers.  These receivers, which are inherently broadband (10 kHz to 30 MHz) allow a limited number of users to tune across the bands, allowing reception on frequencies that are not covered by the WebSDR servers.

At present there are six of these receivers on site:  Three are connected to the TCI-530 Omnidirectional antenna (covering 630-10 meters - 2200 meters is included via a separate E-field whip), two are on the east-pointing log-periodic beam antenna (which overs 40-10 meters) and the newest is connected to the northwest-pointing log-periodic beam antenna (which covers 30-10 meters).

Figure 1: Power supply in a PC case! The PC case housing the power supply was repurposed - because, why not? Click for larger version

Up to this point we had been running 5 KiwiSDRs:  Three of them were powered by a pair of 5 volt, 3 amp linear power supplies that are "dioded-ANDed" together to form a 6 amp power supply and the other two KiwiSDRs were powered from a heavily-filtered 5-volt, 3 amp switching power supply.

In recent months, the dual 3 amp linear supply had become problematic, not being able to handle the load of the three KiwiSDRs, so we had to power down KiwiSDR #3.  With the recent installation of the northwest-pointing log periodic antenna, we were also looking toward installing another KiwiSDR for that antenna and we were clearly out of power supply capacity.

Using an ATX supply as a general-purpose power supply - it's not just the green wire!

If you look around on the Web, you'll see suggestions that you just "ground the green wire" to turn on an ATX supply, at which point you may use it as a general-purpose supply.  While grounding the green wire does turn it on, it's not as simple as that - particularly if you leave the power supply unattended.

For example, what if there is a brief short on the output while you are connecting things, or what if the power browns out (or turns off) for just the "wrong" amount of time.  These sorts of things do happen, and can "trip out" the power supply and it may never restart on its own.

With the site being remote, we couldn't afford for this to happen - so you'll see, below, how we remedied this.

Putting together another power supply:

With six KiwiSDRs, the power supply requirements were thus:

• 5 amps continuous, making the assumption that a KiwiSDR's average current consumption would be about 830 mA - a number with generous overhead.
  

• 9 amps on start-up, presuming that each KiwiSDR would briefly consume 1.5 amps upon power-up, again a value with a bit of overhead.
  

• The power supply must not exhibit a "slow" ramp-up voltage as the KiwiSDRs did not "like" that.

In looking around for a power supply on which to base the design, the obvious choice was an computer-type ATX power supply.  Fortunately, I have on-hand a large number of 240 watt ATX supplies with active power factor correction which are more than capable of supplying the current demands, being rated for up to 22 amps load on the 5 volt supply - more than enough headroom as I would be needing less than half of that, at least with the currently-planned usage.

Circuit description:

Refer to the schematic in Figure 2 for components in the description.

Added filtering:

While these power supplies were already known to be adequately RF-clean (important for a receive site!) from their wide use for the WebSDR servers because we would be conducting the DC outputs outside the box - and to receivers - I felt it important that additional filtering be added.  Having scrapped a number of PC power supplies in the past, I rummaged around in my box of random toroids and found two that had probably come from old PC power supplies, wound with heavy wire consisting of 4 or 5 strands in parallel.  These inductors measured in the area 10s of microHenries, enough for HF filtering when used with additional outboard capacitance.

These filter networks were constructed using old-fashioned phenolic terminal lug strips.  These consist of a row of lugs to which components are soldered - typically with one or two of the lugs used for mounting, and also "grounding".  Rather than mount these lugs using a drill and screw, they were soldered to the steel case itself - something easily done by first sanding a "bare" spot on the case to remove any paint or oxide and then using an acid-core flux - cleaning it up afterwards, of course!

The heavier components (inductors, capacitors) were mechanically secured using RTV (silicone) adhesive to keep them from moving around - and to prevent the possibility of the inductor's wire from touching the case and chafing.

Looking at the schematic you may note that  C202, C302, C501, C502 and C503 are connected to a "different" ground than everything else.  While - at least for this power supply - the "Common" (black) wire is internally connected to the case, it's initially assumed that this lead - which comes from the power supply - may be a bit "noisy" in terms of RF energy, so they are RF bypassed to the case of the power supply.  This may have been an unneeded precaution, but it was done nonetheless.

Connectorizing and wiring the power supply:

The ATX power connector was extracted from a defunct PC motherboard to allow the power supply itself to be replaced in the future if needed.  On this connector, all of the pins corresponding with the 5 volt (red wires), 12 volt (yellow wires) and ground (black wires) were bonded together to form three individual busses and heavy (12 AWG) wires were attached to each:  This was done to put as many of the wires emerging from the power supply in parallel with each other to minimize resistive losses. 

The green wire (the "power" switch) and purple wire (the 5 volt "standby") were brought out separately as they would be used as well - and the remainder of the pins (3.3 volt, -12 volt, -5 volt, "power good", etc.) were flooded with "hot melt" glue to prevent anything from touching anything else that it shouldn't.

The 5 volt supply was split two ways - each going to its own L/C filter network (L501, L502, C502, C503, C504, C505) as shown in the schematic, this being done to reduce the total current through the inductor - both to minimize resistive losses, but also to reduce the magnetic flux in each inductor, something that could reduce its effective inductance.

Although I don't have immediate plans to use the 12 volt supply, a similar filter (L503, C506, C507) was constructed for the 12 volt supply lead.  On the output side of the 12 volt filter, a 3 amp self-resetting thermal fuse (F501) was installed to help limit the current should a fault occur. 

About the self-resetting fuses:

 These fuses - which physically look like capacitors - operate by having a very low resistance when "cold".  When excess current flows, they start to get warm - and if too much current flows, they get quite hot (somewhere above 200F, 100C) and their internal resistance skyrockets, dropping the current to a fraction of its original value:  It's this current flow and their heat that keeps the resistance high.

It's worth noting that these fuses don't "disconnect" the load - they just reduce the current considerably to protect whatever it is connected to it.  Since, when "blown", they are hot, they must be mounted "in the clear" away from nearby objects that could be damaged by the heat - and also to prevent lowering of their trip current by trapping heat or being warmed by another component - such as another such fuse.  

It should be noted that if the outputs - either 5 or 12 volts - are "hard shorted", the thermal fuse may not react quickly enough prevent the power supply from detecting an overcurrent condition and shutting down.  As an output short is not expected to be a "normal" occurrence, this behavior is acceptable - but it will require that the power supply be restarted to recover from shutdown, as described below.

In the case of the KiwiSDRs, they are connected with fairly long leads (about 6 feet, 2 meters) and often have enough internal resistance to reduce the current below the power supply's overcurrent limit and rather than allowing the full current of the power supply (which could be more than 20 amps) to flow through and burn up this cable, the fuse will trip as it should, protecting the circuit.  To "reset" the fuse, the current must be removed completely for long enough for the device to cool - something that is done with the 5 volt supplies as we'll see, below.

The controller:

As mentioned earlier, if you look on the web, you'll see other power supply projects that use an ATX power supply as a benchtop power source and most of those suggest that one simply connect the green (power on) wire to ground to turn it on - but this isn't the whole story.  In testing the power supply, I noticed two conditions in which doing this wouldn't be enough:

• Shorting a power supply output.  If the output of a good-quality ATX power supply is shorted, it will immediately shut down - and stay that way until the mains power is removed (for a minute or so) or the power supply is "shut off" by un-grounding the green wire for a few seconds before reconnecting to "restart" the power supply.
  

• Erratic mains power interruption.  It was also observed that if the mains power was removed for just the right amount of time, the power supply would also shut down and would not restart on its own.  It took the same efforts as recovering from an output short to restart the power supply.

Since this power supply would be at the WebSDR site - an unmanned location in rural, northern Utah - it would require additional circuitry to make this power supply usable.

Fortunately, an ATX power supply has a second built-in power supply that is independent of the main one - the "standby" power supply.  This is a low-power 5 volt supply that is unaffected by what happens to the main supply (e.g. not controlled by the power switch and not affected if it "trips off") and can be used to power a simple microcontroller-based board that can monitor and sequence the start-up of the main power supply.  For this task I chose the PIC16F688, a 14 pin microcontroller with A/D conversion capability and a built-in clock oscillator.

As seen in the schematic, the "5 volt standby" is dioded-ORed (D601, D602) with the main power supply (12 volts) so that it always gets power - from either the 5 volt standby, or from the 12 volt output - when mains is applied.  R603 and capacitor C602 provide a degree of protection to the voltage regulator should some sort of "glitch" appear on the 12 volt supply - possibly due to the 5 volt load being abruptly disconnected (or connected) as the 5 and 12 volt supplies are "co-regulated" in the sense that it's really only the 5 volt output that is being regulated well - the 12 volt power supply's output is pretty much a fixed ratio to the 5 volt and doesn't really have much in terms of separate regulation.

It should be noted that when operating from the standby +5 volt power source, the voltage from U2 (the 5 volt regulator) is on the order of 3 volts or so (drop through D602 and U2) but this is comfortably above the "brownout" threshold of the PIC, which is around 2.5 volts, so there isn't really a worry that the low-voltage brownout detector will trigger erroneously and prevent start-up.  If it had, I would have simply moved the cathode side of D602 to the +5V side of U2.

Figure 3:  Inside the case! Top right:  12 volt supply filtering and thermal fuse Upper-middle:  Dual 5 volt filtering Lower middle:  Controller board with FET switches and thermal fusing. The ATX power supply is in the lower-left corner. Click on the image for a larger version.

• When the microcontroller starts up, it makes sure that the ATX "power on" wire is turned off (e.g. un-grounded).  This is done by the microcontroller turning off Q401.
  

• After a 10 second delay, it turns on the power supply by turning on Q401.
  

It also monitors the power supply to look for a fault.  If either the 5 or 12 volt output is shorted or faults out, both power supply outputs (but not the 5 volt "standby" output) disappear.

• If, while running, the monitored 12 volt supply (via R601/R602 and "12V V_MON") drops below about half the voltage (e.g. trips out) the "power on" wire is turned off using Q401, disabling the ATX power supply.
  

• A 10 second delay is imposed before attempting to turn the power supply back on.
  

• Once the power supply is turned back on, monitoring of the voltage resumes.

In practice, if there is a "hard" short on the output, the power supply will attempt to restart every 10 seconds or so, but remember that a short on an output could occur with ANY sort of power supply, so this isn't a unique condition.

5 volt output sequencing and monitoring:

The other function of the controller is to sequence and monitor the 5 volt outputs.  As mentioned earlier, it was noted that the KiwiSDRs do not "like" a slow voltage ramp-up so a FET switch is employed to effect a rapid turn-on - and since there are two separately-filtered 5 volt busses, there are two such switches.  In order to reduce the peak current caused when the load is suddenly connected, each of these busses is turned on separately, a 10 second delay between the two of them.

The N-channel FET switches (Q203, Q303) are controlled by an NPN (Q201, Q301) transistor being turned on by the microcontroller which, in turns, "pulls" the base of a PNP transistor (Q202, Q302) low via a base resistor (R202/R302), turning it on - and other resistors (R203, R303) assure that these transistors are turned off as needed.

With the emitter of the PNP connected to the 12 volt supply, the gate voltage of the FET is approximately 7 volts higher than the drain voltage, assuring that it is turned on with adequately low resistance.  Capacitors (C201, C301) are connected between the FET's gates and sources to suppress any ringing that might occur when the power is turned on/off and as a degree of protection against source-gate voltage spikes while the 47k resistor (R207/R307) assure that the FET gets turned off.

The use of P-channel FETs was considered, but unless special "logic level" threshold devices were used, having only 5 volts between the gate and drain wouldn't have turned them fully "on" unless the -5 or -12 volt supply from the power supply was also used.  While this would certainly have been practical, N-channel FETs are more commonly available.

Figure 2:  Schematic of the ATX controller with power supply filtering, voltage monitoring, and control. See the text for a description. Click on the image for a larger version.

In series with the 5 volt supply and the FET's source is a 5 amp self-resetting thermal fuse to limit current.  Should an overload (more than 5-ish amps) occur on the output bus, this fuse will heat up and go to high resistance, causing the output voltage to drop.  If this occurs, the microcontroller, which is using its A/D converter to look at the voltage divider on the outputs (R205/R206 for the "A" channel, R305/R306 for the "B" channel) will detect this dip in voltage and immediately turn off the associated FET.  After a wait of at least 10 seconds - for the fault to be cleared (in the event that it is momentary) and to allow the thermal fuse to cool off and reset - the power will be reconnected.  If there continues to be a fault, the reset time is lengthened (up to about 100 seconds) between restart attempts.

Finally, the status of the power supply is indicated by a 2-lead dual-color (red/green) LED (LED701) mounted to be visible from the front panel.  During power supply start-up, it flashes red, during the time delay to turn on the power supplies it is yellow, when operation is normal it is green - and if there is a fault, it is red.  Optionally, another LED (LED702) can be mounted to be visible:  This LED is driven with the algorithm that causes it to "breath" (fade on and off - and on, and off...) to indicate that "something" was working.  I simply ran out of time, so I didn't install it.

* * *

This power supply was put together fairly quickly, so I didn't take as many pictures as I usually would - and I omitted taking pictures of the back panel where the power supply connections are made.  Perhaps it's just as well as while I used a good-quality screw-type barrier strip, it was mounted to a small piece of 1/4" (6mm) thick plywood that was epoxied into the rectangular hole where one would connect peripherals to the motherboard.

As you would expect, the terminals are color-coded (using "Sharpies" on the wood!) and appropriately labeled.  While not pretty, it's functional!

(Comment:  The photo in Figure 3 was taken before I added the circuit to control the "Power On" wire (e.g. Q401) and the diode-OR power (D601, D602) - and it shows the dual-color LED on the board during testing.)

If you are interested in the PIC's code, drop me a note.

This page stolen from ka7oei.blogspot.com

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Figure 1: The completed "Word Metronome".  There are two recessed buttons on the front and the lights on on the left side. Click on the image for a larger version.

Thus, he asked me to make a "word metronome" - a stand-alone device that would provide a visual cue for speaking cadence.  The idea wasn't to make the speech robotic and staccato in its nature, but rather providing a mental cue to provide pacing - something that is always a concern when trying to make a given amount of material fit in a specific time window:  You don't want to go too fast - and you certainly don't want to be too slow and run over the desired time and, of course, you don't want to randomly change your rate of speech over time - unless there's a dramatic or context-sensitive reason to do so.

To be sure, there are likely phone apps to do this, but I tend to think of a phone as a general-purpose device, not super-well suited for most of the things done with it, so a purpose-built, simple-to-operate device with visual indicators on its side that could just sit on a shelf or desk (rather than a phone, which would have to be propped up) couldn't be beat in terms of ease-of-use.

Circuitry:

The schematic of the Word Metronome is depicted in Figure 2, below:

Figure 2: Schematic of the "Word Metronome" (As noted in the text, the LiIon "cell protection" board is not included in the drawing). Click on the image for a larger version.

This device was built around the PIC16F688, a 14 pin device with a built-in oscillator.  This oscillator isn't super-accurate - probably within +/-3% or so - but it's plenty good for this application.

One of the complications of this circuit is that of the LEDs:  Of the five LEDs, three of them are of the silicon nitride "blue-green" type (which includes "white" LEDs) and the other two are high-brightness red and yellow - and this mix of LED types poses a problem:  How does one maintain consistent brightness over varying voltage.

As seen in Figure 3, below, this unit is powered by a single lithium-ion cell, which can have a voltage ranging from 4.2 volts while on the charger to less than 3 volts when it is (mostly) discharged.  What this means is that the range of voltage - at least for the silicon nitride types of LEDs - can range from "more than enough to light it" to "being so dim that you may need to strike a match to see if it's on".  For the red and yellow LEDs, which need only a bit above two volts, this isn't quite the issue, but if one used a simple dropping resistor, the LED brightness would change dramatically over the range of voltages available from the battery during its discharge curve.

As one of the goals of this device was to have the LEDs be both of consistent brightness - and to be dimmable -  a different approach was required - and this required several bits of circuity and a bit of attention to detail in the programming.

The Charge Pump:

Perhaps the most obvious feature of this circuit is the "Charge Pump".  Popularized by the well-known ICL7660 and its many (many!) clones, this type of circuit may also be driven by a microcontroller and implemented using common parts.  Like its hardware equivalent, it uses a "flying capacitor" to step up the voltage - specifically, that surrounding Q1 and Q2.  In software - at a rate of several kHz - a pulse train is created, and its operation is thus:

• Let is start by assuming that pin RC4 is set high (which turns off Q1) and pin RA4 is set low (which turns off Q2.)
  
• Pin RA4 is set high, turning on Q2, which drags the negative side of capacitor C2 to ground.  This capacitor is charged to nearly the power supply voltage (minus the "diode drop") via D1 when this happens.
• Pin RA4 is then set low and Q2 is turned off.
• At this point nothing else is done for a brief moment, allowing both transistors to turn themselves off.  This very brief pause is necessary as pulling RC4 low the instant RA4 is set low would result in both Q1 and Q2 being on for an instant, causing "shoot through" - a condition where the power supply is momentarily shorted out when both transistors are on, resulting in a loss of efficiency.  This "pause" need only be a few hundred nanoseconds, so waiting for a few instruction cycles to go by in the processor is enough.
  
• After just a brief moment pin RC4 is pulled low, turning on Q1, which then drags the negative side of C2 high.  When this happens the positive side of C2 - which already has (approximately) the power supply voltage is listed to a potential well above that of the power supply voltage.
• This higher voltage flows through diode D3 and charges capacitor C4, which acts as a reservoir:  This voltage on the positive side of C4 is now a volt or so less than twice the battery voltage.
• Pin RC4 is then pulled high, turning of Q1.
• There is a brief pause, as described above to prevent "shoot through", before we set RA4 high and turn Q2 on for the next cycle.

It is by this method that we generate a voltage several volts higher than that of the battery voltage, and this gives us a bit of "headroom" in our control of the LED current - and thus the brightness.

Current limiter:

Transistors Q3 and Q4 form a very simple current limiter:  In this case it is "upside-down" from the more familiar configuration as it uses PNP transistors - something that I did for no particular reason as the NPN configuration would have been just fine.

Figure 3: Inside the "Word Metronome".  The 18650 LiIon cell is on the right - a cast-off from an old computer battery pack.  The buttons on the board are in parallel with those on the case and were used during initial construction/debugging. Click on the image for a larger version.

This circuit works by monitoring the voltage across R3:  If this voltage exceeds the turn-on threshold of Q3 - around 0.6 volts - it will turn on, and when this does it pulls the base voltage, provided by R5, toward Q4's emitter, turning off Q3.  By this action, the current will actually come to equilibrium at that which results in about 0.6 volts across R3 - and in this case, Ohm's law tells us that 0.6 volts across 47 ohms implies (0.6/47=0.0128 amps) around 13 milliamps:  At room temperature, this current was measured to be  a bit above 14 milliamps - very close to that predicted.

With this current being limited, the voltage of the power supply has very little effect on the current - in this case, that through the LEDs which means that it didn't matter whether the LED was of the 2 or 3 volt type, or the state-of-of charge of the battery:  The most that could ever flow through an LED no matter what was 14 milliamps.

With the current fixed in this manner, brightness could be adjusted using PWM (Pulse Width Modulation) techniques.  In this method, the duty cycle ("On" time) of the LED is varied to adjust the brightness.  If the duty cycle is 100% (on all of the time) the LED will be at maximum brightness, but if the duty cycle is 50% (on half of the time) the LED will be at half-brightness - and so-on.  Because the current is held constant, no matter what by the current limiter circuit, we know that the only think that affects brightness of the LED is the duty cycle.

LED multiplexing:

The final aspect of the LED drive circuitry is the fact that the LEDs are all connected in parallel, with transistors Q5-Q9 being used to turn them on.  When wiring LEDs in parallel, one must make absolutely sure that each LED is of the exact-same type or else that with the lowest voltage will consume the most current.

In this case, we definitely do NOT have same-type of LEDs (they are ALL different from each other) which means that if we were to turn on two LEDs at once, it's likely that only one of them would illuminate:  That would certainly be the case if, say, the red and blue LEDs would turn on:  With the red's forward voltage being in the 2.5 volt area, the voltage would be too low for the green, blue or white to even light up.

What this means is that only ONE LED must be turned on at any given instant - but this is fine, considering how the LEDs are used.  The red, yellow or green are intended to be on constantly to indicate the current beat rate (100, 130 or 160 BPM, respectively) with the blue LED being flashed to the beat (and the white LED flashing once-per-minute) - but by blanking the "rate" LED (red, yellow or green) LED when we want to flash the blue or white one, we avoid the problem altogether.

Battery charging:

Not shown in the schematic is the USB battery charging circuit.  Implementing this was very easy:  I just bought some LiIon charger boards from Amazon.  These small circuit boards came with a small USB connector (visible in the video, below) and a chip that controlled both charging and "cell protection" - that is, they would disconnect the cell if the battery voltage got too low (below 2.5-2.7 volts) to protect it.  Since its use is so straightforward - and covered by others - I'm only mentioning it in passing.

Software:

Because of its familiarity to me, I wrote the code for this device in C using the "PICC" compiler by CCS Computer Systems.  As it is my practice, this code was written for the "bare metal" meaning that it interfaces directly with the PIC's built-in peripherals and porting it to other platforms would require a bit of work.

The unit is controlled via two pushbuttons, using the PIC's own pull-up resistors.  One button primarily controls the rate while the other sets the brightness level between several steps, and pressing and holding the rate button will turn it off and on.  When "off", the processor isn't really off, but rather the internal clock is switched to 31 kHz and the charge pump and LED drivers are turned off, reducing the operating current of the processor to a few microamps at most.

Built into the software, there is a timer that, if there is no button press within 90 minutes or so, will cause the unit to automatically power down.  This "auto power off" feature is important as this device makes no noise and it would be very easy to accidentally leave it running.

Below is a short (wordless!) video showing the operation of the "Word Metronome" - enjoy!

This page stolen from ka7oei.blogspot.com

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