In my previous post I touched on my Lifelong Morse Code Journey. This time I'll touch on some of the keys that have helped me on my journey to the WORLD!
As I mentioned in my earlier posts the key below was my first REAL telegraph key I used in my journey. . This key is over 60 years old and has been mounted on presswood, plywood, tables, plastic and currently this piece of granite. I cannot recall the number of QSOs I've hammered with this key but it has seen me through Novice, General, Advanced and Extra code exams. As I look back on my journey, not only is this my first piece of ham radio equipment but this instrument is my most PRIZED piece of ham radio equipment.
When I decided to reacquaint myself with Morse Code, this was the key I used to re-start my journey. In 1983 when I decided to get relicensed as a Novice I purchased a Morse Code Practice Oscillator Kit from Heathkit. I brought this out of mothballs to continue my Morse Code journey after a 30 year absence.
Well, it seems that adage about "It's just like riding a bicycle" is true. I connected my childhood straight key to the Heathkit Code Oscillator and pounded away.
It didn't take long for my mind and appendages to work in unison. Thanks to operators like Lawrence N2VGA who made it easy to return to the Morse Code mode.
Now that I was reacquainted with doing Morse Code, I decided it was time for this Old Dog to learn a new trick. Even though I had been away from Morse Code for 30 years, the mode of CW has always been in the back of my mind. Years ago I had purchased a Ten Tec Ultramatic Electronic Paddles to pair with my Ten Ten Omni 546D, only never to use it. Now fully engaged in Morse Code, it was time to continue my Lifelong Morse Code Journey and unbox this key to unlock the door to a NEW WORLD of operating CW. PADDLES!
This was a difficult transition for me. It took weeks getting the paddles adjusted to my speed and touch. For me, this unit was so sensitive. It was an ALL NEW experience. But I was not going to let this small piece of metal defeat me. Operators like Bill, K4NYM known affectionately as "The POTA Activator"; helped me navigate me through my unmitigated disastrous sending on far too many occasions. But for me, I remembered that as a youth Novice I never let my poor sending stop me from continuing getting on the air to improve my skills. You can practice as much as you like but REAL WORLD experiences are what HELPED ME.
And if that wasn't enough to keep my fingers busy, I had a couple of Bencher paddles laying around collecting dusk. So why not dusk them off and let my fingers tippy toe across their paddles. I started off in low gear while getting the feel of these Bencher paddles, The BY-1 and BY-2.
For me, these paddles had a different feel than the electronic Ten Tec Ultramatic paddles. I did not hesitate to jump on the air and make contacts as much as possible while making adjustments to the paddles for my personal feel. It didn't take long for me to get really comfortable at higher speeds while having a QSO
As I became more proficient in using paddles, I decided to get more involved operating more SKCC Straight Key Century Club events and their SKCC Sked Page to garner QSOs. Since the club advocates the use of straight keys, side-sweepers (Cooties) and bugs, it was time for me to try something else new and different. So I built my own Hacksaw Blade Cootie Key. Talk about something WAY, WAY out of my COMFORT ZONE. This key pushed me in a good way. It made me become more aware of my sending and not running characters together. It was tough going but well worth it.
It took hours of practice and QSOs for me to feel comfortable using the Cootie Key.
Parks on the Air QSO with Rob WC1N
There is no doubt in my mind I would never have been able to take on this next key, If I had not tried the Cootie Key (Side-Sweeper) first. Even after many hours of practice, many adjustments and many QSOS, I'm still not where I'd like to be using this next key. What key? "The Vibroplex Bug"
It boggles my mind how many operates have in my opinion mastered the use of this key at speeds which are way above my present skill level. This is how it all started with me and My Vibroplex Bug.
Finally after many hours of practice and on - air QSO, I felt confidence enough with setting my Bug up for higher speeds like this Parks on the Air QSO with Tom NV4H.
Learning how to use these different keys has helped revitalize My Lifelong Morse Code Journey, A couple of years ago I made a vow not to buy any more radios. However, I NEVER vowed not to buy any more Morse Code Keys.
Here 's what's next on my Christmas list.
VizKey from Jim W6JIM collection of Keys
Santa, I've bene a good boy so far this year. H! Ho!
Fig 1: The Hammond 1590 aluminum case housing the FE-5860A rubidium osc- oscillator and other circuitry - the markings faded by time and heat. Click on the image for a larger version.
Recently I was getting ready for the October 14, 2023 eclipse, so I pulled out my two 10 MHz rubidium frequency references (doesn't everyone have at least one?) as I would need an accurate and (especially) stable frequency reference for transmitting: The details of what, why and how will be discussed in a post to be added in the near future.
The first of these - my Efratom LP-101 - fired up just fine, despite having seen several years of inactivity. After letting it warm up for a few hours I dialed it in against my HP Z3801 GPSDO and was able to get it to hold to better than 5E-11 without difficulty.
My other rubidium frequency reference - the FEI FE-5680A - was another matter: At first, it seemed to power up just fine: I was using my dual-trace oscilloscope, feeding the 'Z3801 into channel 1 and the '5680A into channel 2 and watching the waveforms "slide" past each other - and when they stop moving (or move very, very slow) then you know things are working properly: See Figure 2, below, for an example of this.
That didhappen for the '5680A - but only for a moment: After a few 10s of seconds of the two waveforms being stationary with respect to each other, the waveform of the '5680A suddenly took off and the frequency started "searching" back and forth, reaching only as high as a few Hz below exactly 10 MHz and swinging well over 100 Hz below that.
My first thought was something along the lines of "Drat, the oven oscillator has drifted off frequency..."
Fig 2: Oscillogram showing the GPS reference (red) and the FE-5680A (yellow) 10 MHz signals atop each other. Timing how long it takes for the two waveforms "slide" past each other (e.g. drift one whole cycle) allows long-term frequency measurement and comparison. Click on the image for a larger version.
As it turns out, that was exactly what had happened.
Note:
I've written a bit more about the aforementioned rubidium frequency references, and you can read about them in the links below:
While it is the "physics package" (the tube with the rubidium magic inside) that determines the ultimate frequency (6834683612 Hz, to be precise) it is not the physics package that generates this frequency, but rather another oscillator (or oscillators) that produce energy at that 6.834682612 GHz frequency, inject it into the cavity with the rubidium lamp and detect a slight change in intensity when it crosses the atomic resonance.
In this unit, there is a crystal oscillator that does this, using digital voodoo to produce that magic 6.834682612 GHz signal to divine the hyperfine transition. This oscillator is "ovenized" - which is to say, the crystal and some of the critical components are under a piece of insulating foam, and attached to the crystal itself is a piece of ceramic semiconductor material - a PTC (positive temperature coefficient) thermistor - that acts as a heater: When power is applied, it produces heat - but when it gets to a certain temperature the resistance increases, reducing the current consumption and the thermal input and the temperature eventually stabilizes.
Because we have the rubidium cell itself to determine our "exact" frequency, this oven and crystal oscillator need only be "somewhat" stable intrinsically: It's enough simply to have it "not drift very much" with temperature as small amounts of frequency change can be compensated, so neither the crystal oven - or the crystal contained within - need to be "exact".
Fig 3: The FE-5680A itself, in the lid of the case of the 1590 box to provide heat- sinking. As you can see, I've had this unit open before! Click on the image for a larger version.
What is required is that this oscillator - which is "pullable" (that is, its precise frequency is tuned electronically) must be capable of covering the exact frequency required in its tuning range: If this can't happen, it cannot be "locked" to the comparison circuitry of the rubidium cell.
The give-away was that as the unit warmed up, it did lock, but only briefly: After a brief moment, it suddenly unlocked as the crystal warmed up and drifted low in frequency, beyond the range of the electronic tuning.
Taking the unit apart I quickly spotted the crystal oscillator under the foam and powering it up again, I kept the foam in place and watched it lock - and then unlock again: Lifting the foam, I touched the hot crystal with my finger to draw heat away and the unit briefly re-locked. Monitoring with a test set, I adjusted the variable capacitor next to the crystal and quickly found the point of minimum capacitance (highest frequency) and after replacing the foam, the unit re-locked - and stayed in lock.
Bringing it up to frequency
This particular '5680A is probably about 25 years old - having been a pull from service (likely at a cell phone site) and eventually finding its way onto EvilBay as surplus electronics. Since I've owned it, it's also seen other service - having been used twice in in ground stations used for geostationary satellite service as a stable frequency reference, adding another 3-4 years to its "on" time.
As quartz crystals age, they inevitably change frequency: In general, they tend to drift upwards if they are overdriven and slowly shed material - but this practice is pretty rare these days, so they seem to tend to drift downwards in frequency with normal aging of the crystal and nano-scale changes in the lattice that continue after the quartz is grown and cut: Operating at elevated temperature - as in an oven - tends to accelerate this effect.
By adjusting the trimmer capacitor and noting the instantaneous frequency (e.g. adjusting it mechanically before the slower electronic tuning could take effect) I could see that I was right at the ragged edge of being able to net the crystal oscillator's tuning range with the variable capacitor at its extreme low end, so I needed to raise the natural frequency a bit more.
If you need to lower a crystal's frequency, you have several options:
Place an inductor in series with the crystal. This will
lower the crystal's in-circuit frequency of operation, but since doing
so generally involves physically breaking an electrical connection to
insert a component, this is can be rather awkward to do.
Fig 4: The tip of the screwdriver pointing at the added 2.2uH surface-mount inductor: It's the black-ish component at sort of a diagonal angle, wired across the two crystal leads. Click on the image for a larger version.
Place a capacitor across the crystal. Adding a few 10s of pF of extra capacitance can lower a crystal's frequency by several 10s or hundreds of ppm (parts-per million), depending on the nature of the crystal and the circuit.
Since the electrical "opposite" of a capacitor is an inductor, the above can be reversed if you need to raise the frequency of a crystal:
Insert a capacitor in series with the crystal. This is a very common way to adjust a crystal's frequency - and it may be how this oscillator was constructed. As with the inductor, adding this component - where none existed - would involve breaking a connection to insert the device - not particularly convenient to do.
Place an inductor across the crystal. Typically the inductance required to have an effect will have an impedance of hundreds of ohms at the operating frequency, but this - like the addition of a capacitor across a crystal to lower the frequency - is easier to do since we don't have to cut any circuit board traces.
With either method of tweaking the resonance of the oscillator circuit, you can only go so far: Adding reactance in series or parallel will eventually swamp the crystal itself, potentially making it unreliable in its oscillation - and if that doesn't happen, the "Q" is diminished, potentially reducing the quality of the signal produce and furthermore, taking this to an extreme can reduce the stability overall as it starts to become more temperature sensitive with the added capacitor/inductor than just the crystal, alone.
In theory, I could have placed a smaller fixed capacitor in series with the trimmer capacitor - or used a lower-value capacitor - but I chose, instead, to install a fixed-value surface-mount inductor in parallel with the crystal as it would not require cutting any traces. Prior to doing this I checked to see if there was any circuit voltage across the crystal, but there was none: Had I seen voltage, adding an inductor would have shorted it out and likely caused the oscillator to stop working and I would have either reconsidered adding a series capacitor somewhere or, more likely I would have placed a large-value (1000pF or larger) capacitor in series with the inductor to block the DC.
"Swagging" it, I put a 2.2uH 0805 surface-mount inductor across the crystal and powered up the '5680A and after a 2-3 minute warm-up time, it locked. After it had warmed up for about 8 minutes I briefly interrupted the power and while it worked to re-establish lock I saw the frequency swing nearly 100 Hz below and above the target indicating that it was now more less in the center if its electronic tuning range indicating success! As can be seen from Figure 4, there is likely enough room to have used a small, molded through-hole inductor instead of a surface-mount device.
Fig 5: The crystal is under the round disk (the PTC heater) near the top of the picture and the adjustment capacitor is to the right of the crystal. Click on the image for a larger version.
With a bit of power-cycling and observing the frequency swing while the oscillator was hot, I was able to observing the electronic tuning range and in so-doing, increase the capacitance of the trimmer capacitor very slightly from minimum indicating that I now had at least a little bit of extra adjustment room - but not a lot. Since this worked the first time I didn't try a lower value of inductance (say, 1uH) to further-raise the oscillator frequency, leaving well-enough alone.
Buttoning everything back up and putting it back in its case, everything still worked (always gratifying!) and I let the unit "burn in" for a few hours.
Comparing it to my HP Z8530 GPS Disciplined oscillator via the oscilloscope (see Figure 2) it took about 20 minutes for the phase to "slide" one entire cycle (360 degrees) indicating that the two 10 MHz signal sources are within better than 10E-10 of each other - not too bad for a device that was last adjusted over a decade ago and as seen about 15000 operational hours since!
* * *
Follow-up: A few weeks after this was originally posted I had this rubidium reference with me at the Eclipse event as a "hot standby", its frequency being compared to the LPRO-101 - which was the active, on-the-air unit - using an oscilloscope. This (repaired) unit fired up and locked within 5 minutes at the cool (45F/7C) ambient temperature and remained stable for the several hours that it was powered up.
The SDRPlay RSP2pro (left) and RSP1a receivers (right)
The SDRPlay RSP1a is a popular Software Defined Radio (SDR). This device, connected to and powered by the computer via a USB cable covers from VLF through UHF and low microwave frequencies.
This receiver shares a similar internal architecture of similar devices such as the RTL-SDR dongle and the AirSpy in that an analog frequency converter (mixer) precedes the analog-to-digital converter: In the case of the SDRPlay, the frequency to which the receiver is tuned is (usually) converted to baseband I/Q signals, with the "center" frequency being at zero Hz (DC).1
Note:
For the purposes of this discussion, there is no difference between the RSP1a and some of the other receivers in the product lineup (e.g. RSPDuo, RSPdx and the discontinued RSP1, RSP2 and RSP2pro) in terms of harmonic response across the 2-30 MHz range as they all have about the same 12 MHz and 30 MHz cut-off frequencies on their input filtering - properties that would affect HF reception across the 2-30 MHz range in terms of harmonic response.
By its nature, a frequency mixer is a non-linear device. Ideally, the two frequencies applied to a mixer would yield just two more - the sum and difference. For example, if we applied a 5 MHz signal and a 1 MHz signal to a mixer, it would output both the sum of 6 MHz and the difference of 4 MHz - and this is true, but there's more to the story.
In our example - with a real-world mixer, we will also get additional products - including those related to the harmonics of the local oscillator and the applied signal. Because of this, we will see weaker signals at:
11 MHz (2 * 5 MHz + 1 MHz)
9 MHz (2 * 5 MHz - 1 MHz)
7 MHz (5 MHz + 2 * 1 MHz)
3 MHz (5 MHz - 2 * 1 MHz)
And so on.
Typically, these "other" signals will be quite a bit weaker than the original - but they will still be present, possibly at a high enough level to cause issues such as spurious signals - a problem with both receivers and transmitters. Typically, these are tamed by proper design of the mixer, proper selection of frequencies and careful filtering around the mixer to limit the energy of these "extra" signals.
Note: There will be a response at 5x the center frequency as well, but it is suppressed better than the 3x response by the mixer and - for the 80 meter amateur band and higher - these responses are suppressed reasonably well by the filtering.
SDRPlay's poor harmonic response suppression on 80 meters and below.
ANY receiver will experience spurious responses related to mixing products. Typically, filtering is employed to remove/minimize such responses, but for a wide-bandwidth receiver such an SDR, doing this is complicated by the fact that being able to cover wide swaths of bandwidth would ideally require a large number of overlapping filters.
An example of a radio where this is done - albeit of different architecture - is the Icom IC-7300 which has nine overlapping band-pass filters that cover 160 through 10 meters. While the reasons for the '7300 having many filters has as much to do with its being a "direct sampling" 2 type of SDR, good filtering on the signal path of any type of receiver - SDR or "HDR" (Hardware Defined Radio - or an "old school" analog type) is always a good idea
If this many filters had been implemented on the SDRPlay, there would be enough filtering to prevent a significant harmonic response. In the case of the RSP1a, this was not done - partly to allow 5-8 MHz of continuous coverage without being significantly impacted by the filters in many cases, but more likely it was done due to practical reasons of economics3: There are just three filters used for covering all of the "HF" amateur bands 160 through 10 meters: One that covers up to 2 MHz, another that covers 2-12 MHz and third that covers 12-30 MHz: This information is covered in the RSP1a technical information document (https://www.sdrplay.com/wp-content/uploads/2018/01/RSP1A-Technical-Information-R1P1.pdf )
The sensitivity to harmonics was tested with the RSP1a's local oscillator (but not necessarily the virtual receiver) tuned to 3.7 MHz 4. For reasons likely related to circuit symmetry, it is odd harmonics that will elicit the strongest response which means that it will respond to signals around (3.7 MHz * 3) = 11.1 MHz. "Because math", this spurious response will be inverted spectrally - which is to say that a signal that is 100 kHzabove 11.1 MHz - at 11.2 MHz - will appear 100 kHz below 3.7 MHz at 3.6 MHz. (It's likely that there are also weaker responses at frequencies around 5 times the local oscillator, but these are - for the most part - adequately suppressed by the filtering.)
In other words, the response to spurious signals follow this formula:
Apparent signal = Center frequency + ((Center frequency * 3) - spurious signal) )
Where:
Center frequency = The frequency to which the local oscillator on the RSP is tuned. In the example above, this is 3.7 MHz.
Spurious signal = The frequency of spurious signal which is approximately 3x the center frequency. In the example above, this is 11.2 MHz.
Apparent signal = Lower frequency where signal shows up. In the example above, this is 3.6 MHz.
In our example - a tuned frequency of 3.7 MHz - the 3rd harmonic would be within the passband of the 2-12 MHz filter built into RSP1a meaning that the measured response at 11.2 MHz will reflect the response of the mixer itself, with little effect from the filter as the 2-12 MHz filter won't really affect the 11 MHz signal - and according to the RSP1a documentation (link), this filter really doesn't "kick in" until north of 13 MHz.
In other words, in the area around 80 meters, you will also be able to see the strong SWBC (Shortwave Broadcasting) signals on the 25 meter band around 11 MHz.
How bad is it?
Measurements were taken at a number of frequencies and the amount of attenuation is indicated in the table below. These values are from measurement of a recent-production RSP1a and spot-checking of a second unit using a calibrated signal generator and the "HDSDR" program:
LO Frequency
Measured Attenuation at 3X LO frequency
Attenuation in "S" Units
2.1 MHz
21 dB (@ 6.3 MHz)
3.5
2.5 MHz
21 dB (@ 7.5 MHz)
3.5
3.0 MHz
21 dB (@ 9.0 MHz)
3.5
3.7 MHz
21 dB (@ 11.1 MHz)
3.5
4.1 MHz
23 dB (@ 12.3 MHz)
3.8
4.5 MHz
30 dB (@ 13.5 MHz)
5
5.0 MHz
39 dB (@ 15.0 MHz)
6.5
5.5 MHz
54 dB (@ 16.5 MHz)
9
6.0 MHz
54 dB (@ 18.0 MHz)
9
6.5 MHz
66 dB (@ 19.5 MHz)
11
12.0 MHz
21 dB (@ 36.0 MHz)
3.5
12.5 MHz
21 dB (@ 37.5 MHz)
3.5
13.5 MHz
22 dB (@ 40.5 MHz)
3.7
14.5 MHz
26 dB (@ 43.5 MHz)
4.3
15.5 MHz
31 dB (@ 46.5 MHz)
5.2
16.5 MHz
35 dB (@ 49.5 MHz)
5.8
17.5 MHz
39 dB (@ 52.5 MHz)
6.5
18.5 MHz
43 dB (@ 55.5 MHz)
7.2
19.5 MHz
46 dB (@ 58.5 MHz)
7.7
20.5 MHz
50 dB (@ 61.5 MHz)
8.3
21.5 MHz
53 dB (@ 64.5 MHz)
8.8
Table 1: Measured 3rd harmonic response of the RSP1a
Interpretation:
In the above chart we see the local oscillator frequency in the left column, the measured attenuation of the 3rd harmonic response (and its frequency) in the center column, and that amount of attenuation expressed in "S" units. Here, an "S" unit is based on the IARU standard (Technical recommendation R.1) of 6 dB per S unit, which is reflected in programs like SDRUNO, HDSDR and many others.
The attenuation of the 3rd harmonic response was measured by first noting the signal level required to obtain a given reading - typically "S-9" near the fundamental frequency - and then observing the level required to obtain that same reading - within +/-1dB - near the 3rd harmonic frequency, using the relationship formula, above.
Below the cutoff frequency of the relevant filter (nominally 12 MHz for receive frequencies in the range of 2 to 12 MHz, nominally 30 MHz for receive frequencies in the range of 12 to 30 MHz) the harmonic response is limited to that of the mixer itself, which is 21 dB.
We can see that on the 2 to 12 MHz segment, the attenuation related to the 3rd harmonic doesn't exceed 40 dB (which is the low end of what I would call "OK, but not great) until one gets above about 5 MHz (which translates to 15 MHz) and it doesn't get to the "goodish" range (50dB or more) until north of about 5.5 MHz which is borne out by the filter response charts published by SDRPlay.
On the 12 to 30 MHz band the filter has practically negligible effect until one gets above about 20 meters, at which point it gets into the "OK, but not great" range by about 18 MHz, and it doesn't really get "goodish" until north of 20.5 MHz. What this means is that strong 6 meter signals may well appear in the 16.5 to 17.5 MHz range as frequency inverted representations.
If there is a relatively strong signal source in the area of the 3rd harmonic response, it will likely appear at the lower receive frequency where the attenuation of the filter is less than 40 dB or so. The severity of this response will, of course, depend on the strength of that signal, the amount of attenuation afforded by the filters at that frequency, and the amount of noise and other signals present in the range of the fundamental frequency response.
Based on the above data, we can deduce the following:
When the RSP1a is tuned between 2 MHz and (below) 12 MHz, it is using its "2-12 MHz" filter. In this range - and below approx. 4 MHz - the 12 MHz cut-off of the filter has negligible effect in reducing 3rd harmonic response.
What this means is that signals from 6-12 MHz will appear more or less unhindered (aside from the 21 dB reduction afforded by the mixer) when the local oscillator of the receiver is tuned between 2 and 4 MHz.
The 3rd harmonic response across 2-4 MHz - which is the 6-12 MHz frequency range - can contain quite a few strong signals and noise sources such as those from shortwave broadcast stations.
When the RSP1a is tuned between 12 MHz and (below) 30 MHz, it is using its "12-30 MHz" filter. Below about 14 MHz, the 30 MHz cut-off of the filter has negligible effect in reducing 3rd harmonic response.
Signals from 36-40 MHz will appear with just 21-26 dB attenuation when tuned in the range of 12-13.5 MHz.
In most cases there are probably few signals in the 36-40 MHz range that are likely to be an issue when tuning in the 12-13.5 MHz range.
80 meter example:
Connecting
the RSP1 to a known-accurate signal generator set to -40dBm, the signal
level at 3.6 MHz was measured: Maintaining the signal level, the
generator was retuned to 11.2 MHz and the resulting signal level was measured to be 21 dB (a bit more than 3 "S" units) lower than that at 3.6 MHz.
What
this means is is that a "20 over S-9" signal at 11.2 MHz will show up
as an S-9 signal at 3.7 MHz, and an S-9 signal at 11.2 MHz will be
around S-6 at 3.7 MHz. In other words, even a "weak-ish" signal at the 3rd harmonic will show up at the lower frequency.
80/60 meter example:
If you run the RSP1a in a wider bandwidth mode, it is possible to simultaneously see and tune a greater frequency range. For example, let us presume that you wish to cover both 80 and 60 meters using a single RSP1. To do this, you could set the center (LO) frequency to 4.5 MHz and set the sample rate to 5.376 MHz and use the 5 MHz band-pass filter built into the RSP1's converter/mixer chip (the Msi001) to prevent in-band aliasing.
In this configuration 20 meter signals will appear at the top of 80 meters owing to the relationship in the formula that we described above. Taking the 20 meter FT-8 subband at 14.074 MHz as an example, we see that:
4.5 MHz + ((4.5 MHz * 3) - 14.074) = 3.926 MHz
In other words, we will see spectrally-inverted representations of 20 meter FT-8 signals around 3.926 MHz, and the rest of the (upper) portion of 20 meters across the rest (lower) portion of 80 meters where USB signals on 20 meters will show up as LSB signals on 80. We know from the chart above that those signals will be attenuated by between 30 and 39 dB (about 5-6 S-units). This might sound like a lot of attenuation, but it means that a "20 over" signal on 20 meters will appear at around S-7 to S-8 on 80 meters - still quite respectable.
More about filtering and harmonic response
While these spurious responses may not be too much of a problem for the casual user, it will be necessary to add additional filtering to allow the RSP1a to function on par with a modern, SDR receiver from one of the major manufacturers.
Unfortunately, the filtering in the RSP1a is not sufficient in the 80 meter case mentioned above as it doesn't have octave filters (or similar) - but what about 60 or 40 meters?
The table above answers this question. In the case of 60 meters - with the receiver tuned to 5.3 MHz - our 3rd harmonic will land on 15.9 MHz. Based on measurements of the receiver the response of signals around 15 MHz - which corresponds to the 19 meter Shortwave Broadcast Band - will be a bit more than 40 dB down from 40 meters with about 20 dB of this being due to the roll-off of the 2-12 MHz filter - but because this frequency range is inhabited by very strong shortwave broadcasters they are likely to still be quite audible around 60 meters.
The situation is a bit better for 40 meters where the 3rd harmonic is around the 15 meter band. There, the 2-12 MHz filter knocks signals down by 50dB or more, putting them about 70dB below the 40 meter response - on par with about any respectable receiver.
What this means is that for amateur bands below 40 meters it is suggested that additional filtering be applied.
The best solution - and recommended for any software-defined radio (or even older "hardware-defined radios") is to have band-pass filter designed for the specific amateur band in question. This will not only significantly attenuate the harmonic response, but it will also reduce the total amount of RF energy entering the receiver, reducing the probability of overload. The obvious down-side is that it will reduce the flexibility of the receiver in that unless you change/remove it, you won't be able to receive signals well outside the filter's design range.
Another possibility is to add a low-pass filter that is designed to cut off signals above the band of interest. For example, if you have a filter that cuts off sharply above 8 MHz, you will be able to tune 80-40 meters and get reasonable attenuation of the 3rd harmonic response across this entire frequency range.
In the case of 160 meters the RSP1a will automatically select the 0-2 MHz low-pass filter and the 3rd harmonic response will be a respectable 50-ish dB down, depending on frequency.
On 20 meters - where the 3rd harmonic is around 42 MHz - the "12-30 MHz" filter will be selected, but the published response of this filter shows that at 42 MHz its attenuation will be quite limited. Practically speaking, it is unlikely that there will be any signals in this frequency range so there being "only" 20-30dB of attenuation is unlikely to cause a problem in most cases, but one should be aware of this.
What can be done:
In short, none of the currently-made SDRPlay receivers - by themselves - will offer very good performance in terms of harmonic rejection between 2 and 5 MHz and it will be particularly bad on the 80 meter band where strong 25 meter SWBC signals can appear: It is interesting that the ARRL review of the RSPdx (Link here) didn't catch this issue.
It is unfortunate that the designers of the SDRPlay receivers did not add at least one additional low-pass filter in the signal path to quash what is a rather strong response in the 2-6 MHz range - particularly on 80 meters, one of the most popular bands. A low-pass filter with a cut-off frequency of 6 MHz (with attenuation becoming significant above 7 MHz) would ameliorate the harmonic response when tuning across this band. This problem is made even worse by the fact that even antennas that aren't particularly resonant at their harmonic responses (e.g. the antenna for 80 meters) will likely do quite a decent job of receiving signals in the 11-12 MHz area.
The only real "fix" for this is to install additional filtering between the SDRPlay receiver and the antenna. If single-band operation is all that is desired, the best choice will be a band-pass filter designed for the frequency range in question5 - but unless you are dedicating the receiver just for that one band, this isn't really desirable unless you can easily switch/bypass the filter when tuning elsewhere.
A more flexible solution would be to use a low-pass filter. As we noted above, the 12 MHz roll-off of the built-in (2-12 MHz) filter just doesn't do much to suppress signals from 20 meters, but if we had a filter that had a sharp cut off beginning, say, at 8 MHz, we could use it for 80, 60 and 40 meters - such a filter is depicted schematically, below:
8 MHz low-pass filter schematic, designed using ELSIE
This filter is pretty easy to build:
Capacitors 1 and 5 each consist of a 100pF and 470pF in parallel
Capacitor 3 consists of a 680pf and 220pF in parallel - although you could probably get away with two 470pF capacitors in parallel in a pinch.
Inductors 2 and 4 consist of 16 turns on a T50-2 (or 18 turns on a T37-2) toroid using small wire - 24-30 AWG is fine
A small scrap of PC board material - about 2"x4" (5x10cm) is more than large enough to accommodate with the capacitors soldered directly to the foil and inductors held aloft by the capacitors. The connectors should be attached to the PC board directly - or with short lengths of coax, keeping the ground (shield) lead length to an absolute minimum to minimize the probability of ground-loop induced noise currents.
If you have access to a NanoVNA it's quite easy to check the performance. If anything, it may be necessary to spread the turns across the toroid or remove one turn - but this design is quite forgiving.
The obvious down-side for this is that if you are tuning all over the HF spectrum (above 7.5-8 MHz in the case of the filter above) you'd have to manually remove or bypass any such filtering when you tuned beyond the range that the added filter would pass.
Footnotes:
The receivers mentioned at the beginning of the article (SDRPlay, AirSpy HF, RTLSDR, etc.) have analog-to-digital converters that cover only a portion of the HF spectrum, using a frequency mixer to convert a range of frequencies from the range of interest to a lower frequency, which is then fed into the converter. Limiting the amount of spectrum being ingested by the receiver - particularly when appropriate filtering is used - can improve performance, reduce cost, and especially reduce the total amount of data, allowing a modest computer (older PC, Raspberry Pi) to be used with it.
A "direct sampling" type of receiver - such as that found in the Icom IC-7300, IC-7610, the KiwiSDR, Red Pitaya and the RX-888 (when used at HF) and others like them simply "inhale" large swaths of spectrum all at once. Because the analog-to-digital converter itself has a limited amount of total RF signal power that it can handle, radios like the Icoms have filtering that allow the passage of only the (relatively) small portion of the HF spectrum around that to which the receiver is tuned, reducing the probability of overload from strong signals on frequencies well away from those of interest. Other direct-sampling receivers such as the KiwiSDR, Red Pitaya and RX-888 do not necessarily have band-specific filtering as they are intended to be able to receive multiple frequencies across the entire HF spectrum at once and as such, much more care is required in implementation to prevent overload/distortion for these devices.
In the case of the (currently-produced) RSP receivers, the filtering varies depending on model: In the case of the RSP1a, it has a band-pass filter that covers 2-12 MHz while other models have used just a 12 MHz low-pass - the former being capable of rejecting AM broadcast band (e.g. mediumwave) signals from the input of the receiver when tuned to HF, and the latter not. Some units additionally have a separate "notch" (reject) filter that is designed to remove just AM broadcast-band signals. The situation described in this article - the reception of signals around 11 MHz when tuned to 80 meters - is related to the fact that the 2-12 MHz filter represents a 6:1 frequency range which means that over the lower portion of this spectrum, the 12 MHz cut-off of this filter cannot possibly remove responses to the third harmonic, hence the issue described here.
If you are using a program like SDRUno it may not be readily apparent to what frequency the receiver's local oscillator is tuned. If set to "Zero IF" mode, the local oscillator will be tuned at the same place as the center of the waterfall display when it is fully zoomed out - typically indicated by a slight line at the "Zero Hz" frequency there there is a slight amount of noise energy. By default, one cannot directly tune the local oscillator ("Zero IF" frequency) in SDRUno. If you use the "HDSDR" program by I2PHD (et al) you can independently tune the local oscillator and the frequency of the virtual receiver.
SDRPlay receivers are currently in use at a number of well known and public WebSDRs around the world as the "acquisition device" (e.g. receiver). In most cases these receivers - because they are used only for specific amateur bands - are preceded by a band-pass filter for the band that they are covering, completely eliminating issue noted in this article. It was during testing at one of these WebSDRs - a receiver on 80 meters that does not (yet) have additional filtering - that signals were noted across the 80 meter band in the middle of the day that should not have been there at all - and these signals were quickly realized to be the result of a harmonic response in the front end. These responses were then verified and quantified using two other RSP1a receivers (of different production runs) and test equipment during the preparation of this article. When it was convenient to do so, a low-pass filter with a cut-off frequency of 7.5 MHz was installed on this receiver, solving the image problems.
Figure 1: The two generic "can" oscillators tested - both having been found in my "box of oscillators". Click on the image for a larger version.
Sometimes one comes across a device with one of those cheap crystal "can" oscillators that is "close" to frequency - but not close enough. Perhaps this device is used in a receiver, or maybe it's used for clock generation or clock recovery. Such oscillator are available on a myriad of frequencies - although too-often not exactly the right one!
What if we want to "nail" this oscillator to an external (perhaps GPS) reference? If this oscillator were variable, this task would be simplified, but finding a "VCXO" (Variable-Control Crystal Oscillator) on the frequency of interest is sometimes not even possible.
What if there were a way to externally lock a bog-standard crystal oscillator to an external source?
To answer this question, I rummaged through my box of crystal oscillators (everyone has such a box, right?) and grabbed two of them: A standard 4 MHz oscillator and a 19.440 MHz oscillator that has an "enable" pin.
Comment:
This article refers to standard, quartz crystal oscillators and not MEMs or "Programmable" oscillators where the internal High-Q resonating element likely has no direct relationship with the synthesis-derived output frequency.
Injection locking
This is what it sounds like: Take a signal source of the desired frequency - typically very close to that of the oscillator that you are trying to nail to frequency - and inject it into the circuitry to lock the two together.
This technique is ancient: It accounts for the fact that a (wobbly) table of pendulum-type metronomes set "close" to the same tempo will eventually synchronize with each other, and it is the very technique used in the days of analog TV to synchronize their vertical and horizontal oscillators to the sync signals from the incoming signal.
It's still used these days, one notable example being the means by which an Icom IC-9700's internal oscillator may be externally locked to an external 49.152 MHz source (see: http://www.leobodnar.com/shop/index.php?main_page=product_info&products_id=352 ) - and this is done by putting a known-stable source of 49.152 MHz "very near" the unit's built-in oscillator.
Injection-locking a discrete-component crystal oscillator is relatively simple: It's sometimes just a matter of placing a wire near the circuitry with the resonant element (e.g. near the crystal or related capacitors) and the light capacitive coupling will cause it to "lock" to the external source - as long as it's "close" to the oscillator's "natural" frequency.
Getting a signal inside the oscillator
Injection locking often needs only a small amount of external signal to be applied to the circuit in question - particularly if it's inserted in the feedback loop of the resonant circuit, but what about a "crystal can" oscillator that is hermetically sealed inside a metal case?
Figure 2: Schematic depiction of power supply rail to get the external signal "into" the can. Click on the image for a larger version.
Because, in many cases, opening the can would compromise the seal of the oscillator and expose the quartz element to air and degrade it, this isn't really an option. Another possibility would be to magnetically couple an external signal into the circuitry, but owing to a combination of its small size and the fact that these devices are typically in ferrous metal cans, this isn't likely to work, either.
So what else can one do to get a sample of our external signal inside?
Power rail injection
The most obvious "input" is via the power supply rail. Fortunately - or unfortunately, depending on how you look at it - these oscillators often have built-in bypass capacitors on their power rails, putting a low-ish impedance on the power supply input - but this impedance isn't zero.
Figure 3: Top - The signal riding on the voltage rail Bottom - The locked output of the oscillator Click on the image for a larger version.
A simple circuit to do this is depicted in Figure 2. The way it works is by decoupling the power supply via L1 and C1 and heavily "modulating" it with the signal to be locked with Q1. For the test circuit seen in Figures 2 and 4, L1 and L2 were 10uH molded chokes, C1 and C2 were 0.1uF capacitors and Q1 was a 2N3904 or similar NPN transistor.
When an external signal is applied to Q1 via C2 (I used +13dBm of RF from a signal generator) Q1 will conduct on the positive excursions of the input waveform, dragging the power supply voltage to the oscillator down with it. With this simple circuit, Q1 has to dissipate quite a bit of power (the current was about 500 mA) and this action results in a fair bit of power dissipation, likely due to the fact that the bypass capacitance within the oscillator is shunting the energy and causing a significant amount of power to be lost.
This circuit has room for improvements - namely, it's likely that one could better-match the collector impedance of Q1 with the (likely) much lower impedance at the V+ terminal of the oscillator - possibly using a simple matching circuit (L/C, transformer, etc.) to drive it more efficiently.
Figure 4: The messy test circuit depicted in Figure 2 used to inject the external into the "can" oscillator via the power pin. Click on the image for a larger version.
Despite its simplicity, with the circuit in Figure 2 shows how I was able to inject an external signal source into the oscillator and, over a relatively narrow frequency range (15 Hz for the 4 MHz oscillator, 60 Hz for the 19.44 MHz oscillator) it could be locked externally.
The oscillogram in Figure 3 shows the resulting waveforms. The top (red) is the AC-coupled power supply rail for the oscillator showing about 2 volts of RF imposed on it while the bottom rail shows the square-ish wave output of the power supply. Using a dual-trace scope, it was easy to spot when the input and output signals were on the same frequency - and locked - as they did not "slide" past each other.
As you might expect, the phase relationship between the two signals will vary a bit, depending whether one is at the low or high frequency end of the lock range and with changes in amplitude, so this - like about any injection-locking scheme - shouldn't be confused with a true "phase lock".
Is the lock range wide enough?
The "gotcha" here is that these are inexpensive oscillators, likely with 50-100 ppm stability/accuracy ratings meaning that they are going to drift like mad with temperature and applied power supply voltage. What this also means is that these oscillators are not likely to be "dead on" frequency, anyway.
To a degree, their frequency can be "tuned" by varying the power supply voltage: A 5-volt rated "can" oscillator will probably work reliably over a 3.5-5.5 volt range, often changing the frequency by a hundred Hz or so: The 19.44 MHz oscillator moved by more than 1.5 kHz across this range, but never getting closer than 2 kHz above its nominal frequency - but this correlates with the often-loose specifications of these devices in terms of frequency accuracy, not to mention temperature!
If your oscillator is "close enough" to the desired frequency at some voltage - and it is otherwise pretty stable, this may be a viable technique, but other than that, it may just be a curiosity. If one chooses an oscillator with better frequency stability/tolerance specifications - like a TCXO - this may be viable, but testing would be required to determine if a TCXO's temperature compensation would even work properly if the power supply voltage were varied/modulated with an external signal.
"Enable" pin injection
Figure 5: Schematic depicting applying an external signal via the "enable" pin. The amplitude of the external signal must have a peak-to-peak voltage that is a significant percentage of the power supply voltage. Click on the image for a larger version.
Many of these "can" oscillators have (or may be ordered with) an "enable" pin which turns them on and off - and unlike the power supply pin, this typically has pretty low parasitic capacitance compared to the V+ pin of the oscillator and it can provide a way "in" for the external frequency reference. Figure 5 shows how this can be done.
For this circuit, resistors Ra and Rb (which may be between 1k and 10k, each) bias the "enable" pin somewhere around the threshold voltage and capacitively couple the signal - in this case, a +13dBm signal from a signal generator which had about 2 volt peak-to-peak swing. If a logic-level signal is available, one can dispense with the bias resistors and the capacitor and drive it directly.
Note that some oscillators have a built in pull-up or pull-down resistor which can affect biasing and the selection of resistors should reflect that: If its specs note that the pin may be left open to enable (or disable) the oscillator, this will certainly be the case. If a pull-up resistor is present, the value of the corresponding external pull-down resistor will have to be experimentally determined, or "Rb" (in Figure 5) may be made variable using a 10k-100k trimmer potentiometer.
The 19.44 MHz oscillator shown in Figure 1 has such an enable pin and by injecting the 2 volt peak-peak signal from the external source into, it will reliably lock over a 900 Hz range. Some degree of locking was noted even if the signal was quite low (around 250 mV peak-peak) but the frequency swing was dramatically reduced. For optimal lock range it's expected that a swing equal to that of the supply rail would be used.
The precise mechanism by which this works is unknown: Does the "enable" pin actually turn the oscillator on and off, does it simply gate the output of the oscillator while it continues to run or is it that this signal gets into the onboard circuitry and couples into the oscillator's feedback loop? I suspect that it is, in most cases, the former as the "enable" pin often reduces power consumption significantly which would explain why it seems to work reasonably well - at least with the oscillators that were tested.
If the oscillator itself is "gated" (e.g. turned on/off) by the "enable" pin, then this is precisely the mechanism that we would want to inject an external signal into the oscillator. In looking at the output waveform, however, I suspect that the answer to this question isn't that simple: If it were simple logic gating one would expect to see the output waveform of the oscillator gated - and mixing - with the external signal once the latter was outside the "lock" range - but this was not the case for the oscillator tested. I suspect that there might be some sort of filtering or debouncing in the gating circuit, but based on the ease by which locking was accomplished using this oscillator, there was clearly enough of the external signal getting into the oscillator portion itself to cause it to lock readily.
As noted previously, while the lock range was about 900 Hz, the oscillator itself was about 2.5 kHz high, anyway, so it could not be brought precisely onto the nominal frequency. Again, it may be possible to do this with a TCXO equipped with an "enable" pin, but testing would be required for any specific oscillator to determine if this is viable.
"Locked" performance
The testing of spectral purity using either of these methods was only cursorily checked by tuning to the output of the oscillator with a general-coverage receiver and feeding the resulting audio into the Spectran program to see a waterfall display. This configuration allows both the absolute frequency and the lock range to be measured with reasonable accuracy.
It can also tell us a little bit about spectral purity: If there was a terrible degradation in phase noise, it would likely show up on the waterfall display - but when solidly locked, no such degradation was visible.
Although it wasn't tested, it's also likely that locking the oscillator - particularly using the "enable" pin - could be used to "clean up" an external oscillator that is somewhat spectrally "dirty" owing to the rather limited lock range and high "Q" of the "can" oscillator. This is most likely useful for higher-frequency components (e.g. those farther away from the carrier than a few kHz) rather than close-in, low-frequency phase noise - a property which the most inexpensive oscillators likely don't have is anything resembling stellar performance, anyway.
Harmonic locking?
One thing that I did not try (because I forgot to do so) was harmonic locking - that is, the injection of a signal that is an integer fraction of the oscillator frequency (e.g. 1 MHz for the 4 MHz oscillator) - perhaps something to try later?
Is this useful for anything?
I had wondered for some time if it would be possible to lock one of these cheap oscillators to an external source and the answer appears to be "yes". Unfortunately, most crystal oscillators have accuracy and temperature stability specifications that cause its natural frequency variance to exceed the likely lock range unless one gets a particularly stable and accurate oscillator.
If one presumes that the oscillator to be "tamed" is good enough then yes, it may be practical to lock it to an external source - particularly via the "enable" pin. In many cases, such oscillators don't have this feature as they need to be active all of the time so it may be necessary to replace it with one that has an "enable" pin - and then one must hope that the replacement will, in fact, be stable/accurate enough and also capable of being locked externally - something that must be tested on the candidate device.
Figure 1: The NDK 9200Q7 OCXO. This unit, pulled from used equipment, is slightly "shop-worn" but still serviceable. The multi-turn tuning potentiometer is accessible via the hole at the lower-left. Click on the image for a larger version
The NDK 9200Q7 (pictured) is an OCXO (Oven-Controlled Crystal Oscillator) that occasionally appears on EvilBay or surplus sites. While not quite as good a performer as the Isotemp 134-10 (see the 17 October, 2017 Blog entry, "A 10 MHz OCXO" - Link) it's been used for a few projects requiring good frequency stability, including:
The 146.620 Simulcast repeater system. One of these is used at each transmitter site, which are held at 4 Hz apart to eliminated "standing nulls" - and they have stayed put in frequency for over a decade. (This system is described in a series of previous blog entries starting with "Two Repeaters, One System - Part 1" - Link).
10 GHz transverter frequency reference.One of the local amateurs used one of these units to hold his 10 GHz frequency stable and it did so fairly well, easily keeping it within a hundred Hz or so of other stations: This was good enough to allow him to be easily found and tuned in, even when signals were weak.
At least some of these units were pulled from scrapped VSAT (Very Small Aperture SATellite) terminals so they were designed for both stability and the ability to be electronically tuned to "dial in" the frequency precisely.
Testing and experience shows that given 10-15 minutes to thermally stabilize, these units are perfectly capable of holding the frequency to better than 1 part in 108 - or about 1 Hz at 100 MHz - and since any of these units that you are likely to find about are likely to be 25-30 years old, the intrinsic aging of the quartz crystal itself is going to be well along its asymptotic curve to zero.
Figure 2: The bottom of the OCXO, annotated to show the various connections. Click on the image for a larger version.
Using this device
In its original application, this device was powered from a 12-15 volt supply, but if you were to apply power and give it 5-15 minutes to warm up, you would probably be disappointed in its accuracy as it would not have any sort of external tuning input to get it anywhere close to its intended frequency.
Because of the need for it to be electrically tuned, this device is actually a VCXO (Voltage-Controlled Crystal Oscillator) as well and as such, it has a "Tune" pin, identified in Figure 2. Nominally, the tuning voltage was probably between 0 and 10 volts, but unless a voltage is applied, this pin will naturally drift close to zero voltage, the result being that at 10 MHz, it may be a dozen or two Hz low in frequency.
Adding a resistor
The easiest "fix" for this - to make it operate "stand-alone" - is to apply a voltage on the pin. If your plans include locking this to an external source - such as making your own GPSDO (GPS Disciplined Oscillator) then one simply need apply this tuning voltage from a DAC (Digital-to-Analog Converter) or filtered PWM output, but if you wish to use this oscillator in a stand-alone configuration - or even as an externally-tuned oscillator, a bit of modification is in order.
Figure 3: This shows the 10k resistor added between the internal 5 volt source and the "TUNE" pin to allow "standalone" operation. Click on the image for a larger version.
The OCXO may be disassembled easily by removing the small screw on each side and carefully un-sticking the circuit board from the insulation inside. Once this is done, you'll see that there are two boards: The one on the top is part of the control board for the heater/oven while the bottom houses some of the oscillator components.
Contained within the OCXO is a 78L05 five-volt regulator which is used to provide a voltage reference for the oven and also likely used as a stable source of power for the oscillator - and we can use this to our advantage rather than need to regulate an external source which, itself, is going to be prone to thermal changes.
Figure 3 shows the addition of a single 10k resistor on the top board, connecting the "TUNE" pin to the output of this 5 volt regulator. By adding this resistor, the TUNE pin allows one to use this OCXO in a "standalone" configuration with no connection to the "TUNE" pin as it is is automatically biased to a temperature-stable (after warm-up) internal voltage reference and can then be used as-is as a good 10 MHz reference, using the onboard multi-turn potentiometer to precisely set the frequency of operation.
Figure 4: More pictures from inside the OCXO Click on the image for a larger version.
Another advantage of adding the internal 10k resistor is that it's easy to reduce the TUNE sensitivity from an external voltage: This value isn't critical, with anything from 1k to 100k likely being usable. Testing shows that by itself, the oscillator is quite table and varying the TUNE voltage will adjust it by well over 10 Hz above and below 10 MHz.
The inclusion of the 10k internal resistor may also be of benefit. In many cases, having a much narrower electronic tuning range than this will suffice so a resistor of 100k (or greater) can be used in series with the TUNE pin, between it and an external tuning voltage, acting as a voltage divider. Doing this will reduce the tuning range and it can also improve overall stability since much of the tuning voltage will be based on the oscillator's already-stable 5 volt internal source. The stability of the OCXO itself is such that even with a 10-ish:1 reduced tuning range due to a series 100k resistor, there is still far more external adjustment range than really necessary to tune the OCXO and handle a wide range of external temperatures.
The actual value of the added internal resistor is unimportant and could be selected for the desired tuning/voltage ratio based on the external series tuning resistor and the impedance of the tuning voltage.
When reassembling the OCXO, take care that the insulation inside the can is as it was at the time of disassembly to maximize thermal stability and, of course, be sure that the hole in the can lines up with the multi-turn potentiometer!
Operating conditions
Figure 5: Even more pictures from inside the OCXO. Click on the image for a larger version.
The "official" specifications of this OCXO are unknown, but long-term use has shown that it will operate nicely from 12-15 volts - and it will even operate from a 10 volt supply, although the reduced heater power at 10 volts causes warm-up to take longer and there may not be sufficient thermal input for the oven to maintain temperature at extremely low (<15F, <-9C) temperatures unless extra insulation is added (e.g. foam around the metal case.)
It is recommended that if one uses it stand-alone, the voltage source for this device be regulated: While the on-board 5 volt regulator provides a stable reference without regard to the supply voltage, the amount of thermal input from the oven will change with voltage: More power and faster heating at higher voltage. While you might think that this wouldn't affect a closed-loop system, it actually does owing to internal thermal resistance and the fact that due to loss to the environment, there will always be a thermal gradient between the heater, the temperature-sensitive circuitry, and the outside world - and changing the operating voltage and thus the amount of heater power will subtly affect the frequency.
Finally, this oscillator - like any quartz crystal oscillator that you are likely to find - is slightly affected by gravity: Changing orientation (e.g. turning sideways, upside-down, etc.) of this oscillator affects its absolute frequency by a few parts in 10E-8, so if you are interested in the absolute accuracy and stability, it's best to do the fine-tuning adjustment with it oriented in the same way that it will be used and keep it in that orientation.
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