Comment:

This article - while it centers about the investigation of a SolarEdge PV (PhotoVoltaic) system - the discussions of techniques and strategies should be generally useful when investigating interference from any make or model of PV system - or even interference from other sources.

* * *

Several months ago I got a call from a local amateur who was very concerned about a sudden rise in his noise floor across the HF spectrum (3-30 MHz).  This increase in noise seemed to be coincident with the installation and commission of a 5 kW PV (Photovoltaic, or "Solar") electrical system on the house of an adjacent neighbor.  I suggested that he talk to the manufacturer of the PV system to discuss the situation - and to request from them possible solutions.

A few weeks ago, he got back to me and he had, in fact, talked to the manufacturer and an online meeting was arranged in which they would remotely idle the neighbor's system while we were monitoring via the amateur's receive antenna.

Out of curiosity - and as sort of a practice run - I went over the weekend before the online meeting to get a better idea as to the spectral signature of this system - a SolarEdge series string system with optimizers - when it was operating normally.  The amateur had obtained permission from the neighbor to allow us to enter their yard to make very "close in" measurements (e.g. within a few inches/cm of the equipment, conductors) to obtain spectral "samples" of the system, thereby excluding external signals.

For these measurements, I used an amplified, shielded magnetic ("H") loop antenna (about 18"/50cm) in diameter as the "sense" antenna and an HP/Agilent/Keysight spectrum analyzer, recording the plots electronically - although a Tiny SA "Ultra" would likely have sufficed as well.  None of the readings were to represent "absolute" signal levels as all that we were really interested in were relative measurements, and as such all that we needed to do was keep our measurements consistent - that is, being able to precisely repeat the conditions between subsequent measurements.  These readings would allow us to understand the nature of the RF energy that it was creating - its "signature", if you will.

Note:  For information about the H-field loop used for this testing - and using an inexpensive spectrum analyzer such as the "Tiny SA" or, better yet, the "Tiny SA Ultra", see a previous blog entry "RFI Sleuthing with the Tiny SA" - Link.

The nature of this QRM:

The interference observed by this amateur - now known for certain to be signature of this model of SolarEdge PV system - was evident as two general types of signals:

• Moderate to strong clusters of carriers every "even" 200 kHz.  At 200 kHz intervals (e.g. 7.0, 7.2, 7.4 and 14.0, 14.2 and 14.4 MHz) from below 3 MHz through at least 30 MHz could be heard a melange of closely-spaced carriers within about 500 Hz of each other on the lower bands.  While these carriers sounded like mostly CW (unmodulated) signals, there was also evidence of low rate data signalling buried in the cacophony as well as additional lower-level carriers.
  

While the appearance of the above interference coincided with the activation of the neighbor's system, that fact that it disappeared at night further pointed to a PV system as the source of the QRM.

"Close-in" measurements

Placing the sense antenna right at the main inverter on the back of the neighbor's house, we wanted to take a snapshot of the spectrum at that location:

Figure 1:  0-30 MHz sampled right at the main inverter

With each horizontal division representing 3 MHz, this 0-30 MHz plot shows a high concentration of noise in the 3-9 MHz area from a location right at the inverter.

Because Figure 1 represents the spectrum at the inverter, we wondered what it would look like at one of the solar panels so we placed the sense antenna right against one of the solar panels:

Figure 2:  0-30 MHz sweep with sense antenna placed next to a solar panel

Figure 2 is in the same frequency and amplitude scale as Figure 1 - but with the "reference level" adjusted by 20 dB to move the trace "up" a bit - and we can see that the spectrum next to the panel looks quite different from that sampled right at the inverter.  This isn't unexpected as Figure 2 would likely represent more of the noise that is emitted from the DC (input) side of the optimizer whereas the spectrum represented in Figure 1 would be more likely to show that of the DC output of the optimizer plus whatever noise was riding on the conductors carrying the DC input and AC output of the main string inverter.

Although it is difficult to be sure, the 0-30 MHz plots taken from a greater distance (10 meters or more) had the general appearance of the noise spectra shown in Figure 2 more than that of Figure 1 leading me to believe that a significant portion of the QRM may be being radiated from the panels themselves rather than just the conductors going from the optimizers  to the main inverter - but certainly, both are likely involved.

Note:  For both of these plots, the RF energy from the PV system was many 10s of dB above the typical background noise floor - in this case, 40-50dB for Figure 1 and at least 30dB for Figure 2 in the area around 7 MHz.

As the 0-30 MHz sweep does not have enough resolution to visualize the narrower 200 kHz signals, the analyzer was readjusted as depicted in Figure 3 - again with the sense antenna next to the panel:

Figure 3:  From near the panel, a "zoomed in" spectral sweep showing narrowband birdies.

In this spectrum plot we can see not only the "white" noise on the floor of the sweep representing the "hummy hiss", but also the much stronger signals every 200 kHz - plus a number of weaker signals in between:  It is these signals that are the most obvious to the casual operator and appear to be unique to a SolarEdge system of this model/type.

On this same day we waited until after sunset - monitoring the groups of carriers at 7.2 MHz and hearing them "flicker" out of existence as it got dark and we re-did the "next to the panel" measurements - this time the spectrum was devoid of the 200 kHz-spaced carriers (they were no longer audible on the amateur receiver, either) and the 0-30 MHz plots were 10s of dB lower than in the daylight. 

Important:  The 2 MHz sweeps in Figures 3-7 use a resolution bandwidth of 10 kHz which is almost exactly 4 times wider than the typical SSB bandwidth of an amateur receiver of about 2.5 kHz making their apparent level above the background noise appear lower than it is actually is.

What this means is the coherent signals - such as the 200 kHz carriers - appear to be another 6 dB farther above the noise floor in an SSB bandwidth than what the analyzer plots show.

Plots from a distance

Having captured some "close-in" plots, we now had an idea as to what the signals emitted by the PV system looked like.

A few days after we made the above plots we were in a virtual meeting with the manufacturer of the PV system (SolarEdge) from the ham's shack.  Having reconfigured the feed to his main radio, we could quickly switch the feedline from the antenna feeding the radio and the spectrum analyzer.

At this time we also learned that there was a second SolarEdge system south of this amateur's QTH - about 150 feet (45 meters) away across the cul de sac - and that the neighboring system and the one across the street would but remotely shut down, in that order, to determine how much QRM was emanating from each.

While we captured 0-30 MHz plots of each stage of system shutdown, for the purposes of this article we'll show just the "narrow" plots in 2 MHz sweeps as depicted in Figure 3 as the presence of the 200 kHz signal are generally representative of the presence of the broadband noise as well and these signals were easily identifiable and now known to be indicators of QRM from this type of PV system.

First, here's the plot from the amateur's 40 meter inverted Vee antenna with both systems on:

Figure 4:  6-8 MHz plot from the 40 meter antenna showing the 200 kHz peaks - and a bit of broadband noise as well.

 

The next plot shows the effects when the neighboring system was turned off, but the one across the street still on:

Figure 5:  The neighboring system off - but the one across the street still on.

As can be seen, the broadband noise floor around the 40 meter band (approximately one horizontal division below and above the marker) has dropped visibly - around 3-4 dB - and the amplitude of the carrier at 7.2 MHz has dropped about 6 dB - and the 200 kHz signals have disappeared almost entirely below about 6.5 MHz.  The system across the street was then shut off and the only remaining signals were those that happened to be on the 40 meter band.  (No trace is available for this configuration, unfortunately.)

As the 40 meter inverted Vee is oriented to favor east-west signals it was not necessarily the best candidate to test the effects of the PV system across the street, so we switched to a 20 meter antenna which was responsive in that direction and this trace shows the plot between 13 and 15 MHz:

Figure 6:  This plot of the 20 meter band and surrounding frequencies shows only propagated signals, with no sign of PV system QRM.

As both systems were off, the trace was quite clear - only showing signals that actually were on or near the 20 meter band, propagated from elsewhere in the world.  The folks at SolarEdge then turned on the system across the street with the following result:

Figure 7:  Same as Figure 6, but with only the PV system across the street activated.

The effect is clear:  In the vicinity of the 20 meter band, the appearance of rather strong signals every 200 kHz is apparent - and there is an obvious 2-4 dB increase in the noise floor indicating that this system, too, is causing harmful interference.

Readings on the radio:

It would seem that the folks at SolarEdge had worked with more than one amateur radio operator on similar issues and I was pleasantly surprised when they asked for some "S-Meter" reading comparisons with the neighbor's system on and off.  Using a calibrated signal generator, I'd already determined the signal level (in dBm) that correlated with the S-meter readings for the Icom radio - and here are the results for 40 meters:

Both systems off:

S1 (<= -84 dBm) - no carrier groups every 200 kHz.

Neighbor system on:

S4 (-78 dBm) - white noise between 200 kHz carriers.

S9 (-67 dBm) - carriers at 7.2 MHz.

This shows that at 40 meters, the degradation to noise alone was on the order of 6 dB (most Japanese radios are calibrated for 3dB per S-unit) and that the cluster of carriers on 200 kHz intervals was far more destructive, rising a bit short of 20dB out of the noise floor.

As our time with the SolarEdge folks in the virtual meeting was limited, we were not able to do similar "S-meter" tests on 20 meters, but we can use the 40 meter results along with the relative strength of the 200 kHz-spacing carriers  and correlate them with the 40 and 20 meter spectrum analyzer traces and determine that the severity of QRM from the PV system on 20 meters on the receiver would have been roughly comparable to that on 40.

Analysis of these readings and implications:

As mentioned earlier, there are two types of interfering signals produced by these SolarEdge PV systems:

• Moderate to strong clusters of carriers every "even" 200 kHz.  These are very obvious, easy to identify, and quite strong compared to the noise with a few weaker signals in-between that were also clearly audible.
  

Gisteravond meegedaan voor de tweede keer aan de NTC (Netherlands Telegraphy Club) QSO Party (NTCQP). Een morse code / CW party waarin je een uur lang contacten legt met andere stations en gegevens uitwisselt. In dit geval minimaal een ‘echt’-rapport, je naam en NTC nummer. Meer gegevens mag, hoeft niet. Ik heb 11 QSO’s kunnen loggen als S&P (Searching and Pouncing, dus een station opzoeken en proberen te werken). Ik vind S&P gewoon een stuk relaxter.

RUMlogNG en N1MM+
Ik heb met veel gedoe en omwegen in N1MM+ kunnen loggen en met AnyDesk het beeldscherm van de laptop overgenomen op mijn iMac. Ik wilde namelijk voor de NTCQP het logboek programma N1MM+ gebruiken op de laptop en dan koppelen via het netwerk aan RUMlogNG voor de frequentie.

In RUMlogNG is deze QSO Party niet beschikbaar en in N1MM+ wel. Gelukkig zit er in RL een mogelijkheid om RL met N1MM+ te laten praten maar ik kreeg het niet voor elkaar. Ik kreeg van de RUMlogNG auteur als reactie: als je RL niet wilt gebruiken moet je de TRX aan de laptop hangen“. Lekker dan. Dus dat uiteindelijk gedaan en de TRX ging spontaan op zenden…… en dat kreeg ik eerst niet goed. Snel vermogen minimaal (4 Watt) en een paar keer hard aan / uit. Uiteindelijk werkte het en zonder dat ik het in de gaten had ging N1MM+ ineens CQ roepen 😱😱 man man man….

Uiteraard werd ik door het RBN opgepikt waarna Rob HA7RJA op mijn CQ reageerde. Helaas hoorde ik Rob niet want ik had het geluid van de TRX heel zacht staan en was veel te druk met het uitzoeken van het hoe en waarom haha. Sry Rob 😉

Gemak dient de HAM tijdens de contest
Je hoeft eigenlijk alleen maar de call van het tegenstation te ‘nemen’ en in te voeren in N1MM+ en daarna een paar keer klikken (als je dit zou willen dan). Is wel erg makkelijk zo. Daarom maken ze 60! QSOs in een uur. Word je ook niet moe van. Ik heb verder alle communicatie uitgeschakeld, alleen de band, mode en frequentie kloppen nu. De rest doe ik met het handje. Dus seinen met de paddle.

En ja, de naam en het NTC nummer worden dan wel automatisch ingevuld. Ook dat kan ik uitschakelen. Maar heb dat laten staan. Ter controle moet je het toch meelezen.

Storing QRM
Ik heb alleen meegedaan op 40 meter, ondanks de vervelende storing van S9 rond 7.036 MHz en die zwabbert wat heen en weer. Gelukkig zijn CW signalen smalbandig en lukte het aardig door de redelijk goede condities. .

Deze storing is op 80 meter helemaal heftig (S9 + 15dB) met veel QRM tussen de pieken, dus daar heb ik geen moeite gedaan om QSO’s te maken.

Ik had na de vorige NTCQP met mijzelf afgesproken om dit keer meer QSO’s te maken dan de eerste NTC QSO Party. En ja, dat is dus ruim gelukt. Voor de volgende keer wil ik wederom van mijzelf winnen. Dus dan minimaal 12 QSO’s in het log. De volgende NTC QSO Party is op 18 april 2024.

Het bericht NTC QSO Party maart 2024 verscheen eerst op PE2V.

Gisteren weer even meegedaan met de ICWC MST Medium Speed Contest als S&P (Searching and Pouncing, dus een station opzoeken en proberen te werken).

Deze contest wordt georganiseerd door de International CW Council. Bij deze contest ligt de snelheid tussen de 20 en 25 woorden per minuut. De uitwissing is naam en volgnummer.

De condities waren erg slecht en in combinatie met de helaas toegenomen QRM / storing heb ik maar 5 stations kunnen loggen, waarvan enkel 1 uit Amerika, wat ook nog eens moeilijk ging.

Ik hoop dat de condities niet nog slechter worden want naast de K1USN SST (Slow Speed (con)Test en de FISTS Ladder is de MST een leuke contest om aan mee te doen. Het grote voordeel is dat het maar een uurtje duurt, dus lichamelijk prima vol te houden, iets dat helaas wel belangrijk is voor mij.

Verder weinig CW (morse code) verbindingen gemaakt. Ik heb het gevoel dat ik toch de snelheid wat kwijtraak nu de CWops CW Academy Class is gestopt. Dat merk ik vooral bij de oefeningen op lcwo. Vanaf nu toch iedere dag een uurtje morse gaan oefenen want het weg laten zakken is geen onderdeel van mijn CW / morse code missie haha

Het bericht MST Medium Speed conTest (18 mrt 2024) verscheen eerst op PE2V.

One of the (many) banes of the amateur radio operator's existence is often found at the end of an Ethernet cable - specifically a device that is being powered via "Ethernet":  It is often the case that interference - from HF through UHF - emanates from such devices.

Figure 1: POE camera with both snap-on ferrites installed - including one as close to the camera as possible - and other snap-on/toroids to suppress HF through VHF. Click on the image for a larger version.

Why this happens

Ethernet by itself is usually relatively quiet from an (HF) RF standpoint:  The base frequency of modern 100 Megabit and gigabit Ethernet is typically above much of HF and owing to the fact that the data lines are coupled via transformers making them inherently balanced and less prone to radiate.  Were this not the case, the integrity of the data itself would be strongly affected by the adjacent wires within the cable or even if the cable was routed near metallic objects as it would radiate a strong electromagnetic field - and any such coupling would surely affect the signal by causing reflections, attenuation, etc.

This is NOT the case with power that is run via the same (Ethernet) cable.  Typically, this power is sourced by a switching power supply - too often one that is not filtered well - and worse, the device at the far end of the cable (e.g. a camera or WiFi access point - to name two examples) is often built "down to a cost" and itself contains a switching voltage converter with rather poor filtering that is prone to radiation of RF energy over a wide spectrum.  Typically lacking effective common-mode filtering - particularly at HF frequencies (it would add expense and increase bulk) - the effect of RF radiating from the power-conducting wires in an Ethernet cable can be severe.

Even worse than this, Ethernet cables are typically long - often running in walls or ceilings - effectively making them long, wire antennas, capable of radiating (and intercepting) signals even at HF.  The "noisy" power supply at one or both ends of this cable can act as transmitters.

What to do

While some POE configurations convey the DC power on the "spare" conductors in an eight conductor cable (e.g. the blue and brown pairs), some versions use the data pairs themselves (often using center-tapped transformers in the Ethernet PHY) meaning that it may not be easy to filter just the DC power.

While it is theoretically possible to extract the power from the Ethernet cable, filter it and and reinsert it on the cable, the various (different) methods of doing this complicate the matters and doing so - particularly if the DC is carried on the data pairs - can degrade the data integrity by requiring the data to transit two transforms incurring potential signal attenuation, additional reflection and affecting frequency response - to name just a few.  Doing this is complicated by the fact that the method of power conveyance varies as you may not know which method is used by your device(s).

It is possible to subject the entire cable and its conductors to a common-mode inductance to help quash RFI - but this must be done carefully to maintain signal integrity.

Comment: 

Some POE cameras also have a coaxial power jack that permits it to be powered locally rather than needing to use POE.  I've observed that it is often the case that using this local power - which is often 12-24 volts DC (depending on the device) - will greatly reduce the noise/interference generated by the camera and conducted on the cable - provided, of course, that the power supply itself is not a noise source.  Even if a power supply is used near the camera, I would still suggest putting its DC power cable through ferrite devices as described below to further-reduce possible emissions.

There are some devices (such as those sold by DX Engineering) that are essentially back-to-back signal transformers that can reduce radiation of signals from Ethernet cable, but these typically do not permit the passage of power and are not candidates for use with POE devices.

Ferrite can be your friend

For VHF and UHF, simple snap-on ferrites can significantly attenuate the conduction of RF along, but these devices are unlikely to be effective at HF - particularly on the lower bands - as they simply cannot add enough impedance at lower frequencies.

To effectively reduce the conduction of RF energy at HF, one could wrap the Ethernet cable around a ferrite toroidal core, but this is often fraught with peril, particularly with cable carrying Gigabit Ethernet - as tight radius turns can distort the geometry of typical CAT-5/6 cable, affect the impedance and cause cross-coupling into other wire pairs.  If this happens, one often finds that the Ethernet cable doesn't work reliably at Gigabit speeds anymore (being stuck at 100 or even 10 Megabits/second) or starts to "flap" - switching between different speeds and/or slowing down due to retransmissions on the LAN.

One type of Ethernet cable that is quite resistant to geometric distortion caused by wrapping around a toroidal core is the flat Ethernet cable (sometimes erroneously referred to as "CAT6" or "CAT7").  This cable is available as short jumpers around 6 feet (2 meters) long and, with the aid of a female-female 8P8C (often called "RJ-45") coupler can be inserted into an existing Ethernet cable run - just be sure that it is from a reputable source and rated for "Gig-E" service.  As it is quite forgiving to being wrapped around ferrites, this flat cable can be pre-wound with such devices and inserted at the Ethernet switch end and/or the device end at a later time.  I have found that with reasonable quality cable and couplers that this does not seem to degrade the integrity of the data on the LAN cable - at least for moderate lengths (e.g. 50 feet/15 meters or less) - your mileage may vary with very long cable runs.

As the flat cable and female-female Ethernet coupler are to be inserted into the cable run, they must be of known, good quality so it is best to test the couplers and cable that you obtain prior to installation to be sure that their use doesn't cause a reduction in signal quality/speed.

Practical examples

Best attenuation across HF

Figure 2: Three toroids wound on "flat" Ethernet cable.  An FT114-43 is used on each end with an FT114-31 in the middle. Click on the image for a larger version.

One might be tempted to use the more-available FT-240 size of toroids (2.4", 60mm O.D.) but this is unnecessarily large, comparatively fragile and expensive:  While you can fit more turns on the larger toroid, one hits the "point of diminishing returns" (e.g. little improvement with additional turns) very quickly owing to the nature of the ferrite and coupling between turns.  Using the FT114 or FT140 sizes is the best balance as it may be much less expensive than a larger device, it can accept 6-8 turns with the cable's connector installed, and more than 6-8 turns is rapidly approaching the point of diminishing returns for a single ferrite device, anyway.

In bench testing with a fixture, it was found that three toroids on a piece of flat Ethernet cable provided the best, overall attenuation across HF and to 6 meters - significantly better than any combination of FT114, FT140 or FT240 toroids of either 43 or 31 mix alone:  Figure 2, above, shows what this looks like.  Two FT114-43 and one FT114-31 toroid were used - the #31 toroid being placed in the center, providing the majority of series impedance at low HF and a #43 at each end being more effective at higher HF through 6 meters.

To construct this, the flat Ethernet cable was then marked with a silver marker in the center and four turns were wound from each end, in turn, for a total of eight turns on the FT114-31.  Placing an FT114-43 at 12 inches (25cm) and winding seven turns puts the FT114-43 fairly close to each connector, allowing the installation of one or two snap-on ferrites very close  to the connector if it is determined that more suppression is required to suppress radiation at VHF frequencies.  Small zip-ties (not shown in Figure 2) are used to help keep the turns from bunching up too much and also to prevent the start and stop turns from getting too close to each other:  Do not cinch these ties up enough to distort the geometry of the Ethernet cable as that could impact speed - particularly when using Gig Ethernet.

It is important that, as much as possible, one NOT place a "noisy" cable in a bundle with other cables or to loop it back onto itself - both of which could cause inadvertent coupling of the RFI that you are trying to suppress into the other conductors - or to the far side of the cable you are installing.

Best attenuation at VHF and HF

If you are experiencing interference from HF through VHF, you will need to take a hybrid approach:  The use of appropriate snap-on and toroidal ferrite devices.  While snap-on ferrite devices are not particularly useful for HF - especially below about 20 MHz - they can be quite effective at VHF, which is to be expected as that is the purpose for which they are typically designed.  Similarly, a ferrite toroid such as that described above - particularly with type 43 or 31 material - will likely have little effect on VHF radiation - particularly in the near field.

Figure 3: A combination of a snap-on device with an extra turn looped through it and two ferrites to offer wide-band suppression from HF through VHF. Click on the image for a larger version.

Figure 3 shows such a hybrid approach with a snap-on device on the left and two toroids on the right to better-suppress a wider range of frequencies.  In this case it is important that the snap-on device be placed as close to the interference source as possible (typically the camera) as even short lead lengths can function as effective antennas at VHF/UHF.  You may also notice that the snap-on has two turns through its center as this greatly improves efficacy at medium/low VHF frequencies but may be counter-productive at high VHF/UHF frequencies owing to coupling between turns.

Doing this by itself is not likely to be as effective in reducing radiation at VHF/UHF from the cable itself, often requiring the placement of additional ferrite devices.  Figure 1 shows the installation of several snap-on devices placed as close to the POE camera as physically possible - mainly to reduce radiation at VHF and UHF as at those frequencies where even a few inches or centimeters of cable emerging from the noise-generating device can act as an effective antenna.

Determining efficacy

During the installation of these devices on my POE cameras I was interested in how much attenuation would be afforded at VHF:  Since I'd already used the "chokes on a flat cable" approach like that in Figures 2 and 3 I knew that this would likely be as effective as was practical at HF - but because the VHF/UHF noise could be radiated by comparatively short lengths of "noisy" cable - and that the 43 and 31 mix ferrites were probably not as effective at those frequencies - I needed to be able to quantify that what I did made a difference - or not.

Figure 4: The cable in Figure 3 installed, but not yet tucked into place as depicted in Figure 1. (This does not show the snap-on ferrites installed where the wire exits the camera housing.) The female-female RJ45/8P8C "splice" can be seen in the upper-left corner of the picture. Click on the image for a larger version.

For HF this was quite simple:  I simply tuned my HF receiver - connected to my main antenna - to a frequency where I knew that I could hear the noise from the cameras and compared S-meter readings with the system powered up and powered down.  This approach is best done at a time during which the frequency in question is "dead" or at least weak (e.g. poor propagation) - 80/40 meters during the midday and 15/10 meters at night is typical.

For VHF this required a bit more specialized equipment.  My "Go-To" device for finding VHF signals - including noise - is my VK3YNG DF sniffer which has extremely good sensitivity - but it also has an audible "S-meter" in terms of a tone that rises with increasing signal level:  This allowed an "eyes and hands off" approach in determining efficacy of the installation of a ferrite device simply by hearing the pitch of the tone..  Switching it to this mode and placing it and its antenna at a constant distance fairly close to the device being investigated allowed me to "hear" - in the form of a lower-pitched tone - whether or not the application of a ferrite device made a difference.

Slightly less exotic would be an all-mode receiver capable of tuning 2 meters such as the Yaesu FT-817, Icom IC-706, 703 or 705.  In this case the AM mode would be selected and the RF gain control advanced such that the noise amplitude audibly decreased:  This step is important as not doing this could mean that if the noise decreased, the AGC in the receiver would simply compensate and hiding the fact that the signal level changed.  By listening for a decrease in the noise level one can "hear" when installing a snap-on ferrite made a difference - or not.

One cannot use a receiver in FM mode for this as an FM detector is designed to produce the same amount of audio (including noise) at any signal level:  A strong noise source and a weak one will sound exactly the same.  It's also worth noting that the S-meter on a receiver in FM mode - or an FM-only receiver - are typically terrible in the sense that their indications typically start with a very low signal and "peg" the meter at a signal that isn't very strong at all which means that if you try to use one, you'll have to situate the receiver/antenna such that you get a reading that is neither full-scale or at the bottom of the scale to leave room for the indication of change.

Of course, a device like a "Tiny SA" (Spectrum Analyzer) could be used to provide a visual indication, using the "Display Line", markers and stored traces to allow a quick "before and after" determination.  As mentioned above, one would want to place the antenna and the receiving device (an actual receiver or spectrum analyzer) at a fairly close distance to the device being investigated - but keep it and its antenna in precisely the same location (or connected to a fixed-location antenna) during the entire time so that one can get meaningful "before and after" readings.

Conclusion

With the use of ferrites alone, one should not expect to be able to completely suppress radiation of RF noise from an Ethernet cable - the typical maximum to be reasonably expected is on the order of about 20dB (a bit over 3 "S" units) and this can vary wildly with frequency.  In a situation where the POE device is very close to the antenna, it may not be possible to knock the interference down to the point of inaudibility in which case relocation to place the two farther apart - or trying similar devices of different models/brands to try to find a combination to reduce it..

The most effective use will be for noise sources will be at some distance from the receive antenna - particularly if a long cable is used that may act as an antenna.  Additionally, these measures can be effective in situations where your transmitter causes problems with the device itself due to ingress of RF energy along the Ethernet cable.

Be prepared to install appropriate ferrite devices at both ends of the cable as it's often the case that not only does the POE device itself (camera, wireless device) radiates noise but also the POE switch itself:  No-name brand POE power supplies and Ethernet switches are, themselves often very noisy and the proper course of action would be to first swap out the supply or POE switch with a known quiet device before attaching ferrite.

As every interference situation is unique, your mileage may vary, and the best road to success is being able to quantify that changes you have made made things better or worse.

This page stolen from ka7oei.blogspot.com

[END]

DQRM In The 3Y0J Pileups. In 51 years a Ham Radio operator, I've never heard what we just witnessed in almost every Bouvet pileup. We've always had DQRM but...

Figure 1:  Half of the array on my garage - the other half is on the west-facing aspect. There's a bit of shade in the morning around the end of June, but it detracts little during the peak solar production of the day - the hours on either side of "local" noon. Click on the image for a larger version

Personally, I thought that the article was a bit narrow in its scope, with an unsatisfying conclusion (e.g. "The QRM is still there after a lot of effort and expense, but I guess that it's OK") - but this impression is understandable owing to the constraints of the medium (magazine article) and the specific situation faced by the author.

Solar power need not cause QRM:

I can't help but wonder if others that read the article presumed that amateur radio and home solar were incompatible - but I know from personal experience that this is NOT necessarily the case:  There are configurations that will not produce detectable QRM on amateur bands from 160 meters and higher.

Before I continue, let me state a few things important to the context of this article:

Expertise in HF radio interference and home solar installations seems to mutually exclusive - which is to say that you will be hard-pressed to find anyone who is familiar with aspects of both.  This means that in the solar industry itself, you will not likely find anyone who can offer useful advice in putting together a system that will not contribute to the crescendo of electrical noise.

I have heard that many installers (at least in my area) will strongly pressure their potential customers to use microinverter-based systems - and this my experience as well:  From the very start of the process, I was adamant that the design of my system would be series string using SunnyBoy inverters which were known to me to be RF-quiet.  If your installer will not work with you toward your goals, consider a different company.

Designing an "RF-quiet" system as described here may incur a trade-off in available solar production as the use of microinverters can eke out additional efficiencies when faced with issues such as shading and complicated roofs that present a large number of aspects with respect to insolation (e.g. amount of light energy that can be converted to electricity).  Only in the analysis of proposed systems appropriate for your case can you reasonably predict the magnitude of this and whether or not you find it to be acceptable.

What is presented here is my own experience and that of other amateur radio operators with similar PV (PhotoVoltaic) system.  The scope of this experience is necessarily limited owing to the fact that when spending tens of thousands of dollars, one will understandably "play it safe" and pick a known-good configuration.

I will be the first to admit that there are likely other "safe" (low RF noise) combinations of PV equipment that can be demonstrated to be "clean" in terms of radio frequency interference.  I have anecdotally heard of other configurations and systems, but since I have not looked at them first-hand, I am not willing to make any recommendations that could result in the outlay of a large amount of money.  For this reason, please don't ask me a question like "What about inverter model 'X' - does it cause RFI?" as I simply cannot answer from direct experience.

An example system:

The system at my house consists of two series-string Sunny Boy grid-tie inverters:  I can unequivocally state that this system, which has both a SB 5000TL-US-22 (5 kW) and an SB3.8-1SP-US-40 (3.8kW) does not cause any detectable RF interference on any HF frequency or 160 meters - and I have yet to detect any interference on 6 meters, 2 meters or 70cm.  Near the LF and lower MF band (2200 and 630 meters, respectively) some emissions from these inverters can be detected - but none of the switching harmonics (about 16 kHz) land within either of these bands.  

Figure 2:  One of two inverters in the garage.  The Ethernet switch (upper right) produces more RF noise than the inverter! Click on the image for a larger version

Simple roof configuration can equal low noise:

The "simple" roof also has another advantage:  All panels on the faces are oriented the same and a larger number of panels may simply be wired in series.

This simple fact means that known-quiet series-string inverters may be used and known noise-generating components may be omitted from the system - namely, many models of "microinverters" and optimizers.  Both of these devices - despite being very different in their operation - are installed on a "per panel" basis and able to adjust the overall contribution of each panel to maximize the energy input of the entire solar power system.

Having each panel individually optimized for output power sounds like a good idea - and in most cases it is - but this nicety should be taken in context with the goals in mind - but considering that the panels themselves represent a rather small portion of the overall system cost, efficiency losses from not having optimizers can often be offset with the addition of more panels.  To be fair, it is not always possible to simply "add more panels" to make up for loss of production - but this must be carefully weighed against a major goal, which is to produce a "noise free" PV system.

The options have changed:

Since the 2016 article was written, the number of options for series-string inverters has significantly increased and the prices have gone down, allowing options to be considered now that may have been dismissed at that time.  Take the article as an example.

From the photographs accompanying the article, there appear to be two different aspects of panels:  A large array consisting of 30 panels, all seeming to face the same direction;  a smaller array of 8(?) panels:  There appears to be an array of 4 panels, but let us presume that this is an independent energy system.

Assuming that each panel is rated for 300 watts (likely higher than a circa-2016 panel) and that one would wish to limit the maximum open-circuit potential to about 450 volts, this implies the use of at least four MPPT circuits:  The 8 panel array and three arrays consisting of 10 series panels, each.  The maximum output of this system would theoretically be about 11.4 kW - but since one can optimistically expect to attain only about 80% of this value in a typical installation the use of an inverter system capable of 10 kW, as stated in the article, is quite reasonable.

Back in 2016, it would be reasonable to have a 10kW series string inverter with two MPPT inputs representing two separate inputs that could be independently optimized.  If such an inverter were used, this would mean that one input would have just 8 panels and the other would have all 30 panels on the main array - not particularly desirable in terms of balancing.  While all 30 of the panels in the larger array would ostensibly be producing the same output, snow, leaves and shading might cause the loss of efficiency should certain parts be thus impaired.

Having already ruled out the optimizing of each panel independently in the interest of having a "known-quiet" system, we might want to split things up a bit.  As an example, a single 10kW inverter with two MPPT inputs could be replaced with a pair of 5 kW inverters, each with 3 MPPT inputs and having a total of six independent DC inputs allowing the 8 panels of the isolated roof to be optimized together and the remaining 30 panels being divided into 5 arrays of about 6 panels, each.

The 2016 article did not mention the price the system, but a reasonable estimate for that time would be around US$35000 - and it was mentioned, in passing, that the cost of RFI mitigation might have been about 10% of the total system cost, implying about $3500 - about the cost of two Sunny Boy  SB5.0 5 kW series-string inverters, each with three MPPT inputs.

Replicating success:

At least two other local amateur radio operators used the same recipe for low-noise PV systems:  Series-string SunnyBoy grid-tie inverters - specifically the SB 3800TL, SB 5000TL and SB3.8s.  In none of these cases could RFI be detected that could be attributed to the inverter - and the only noise to be detected was with a portable shortwave receiver held within a few inches of the display.

What is known not to be quiet:

From personal experience I know for certain that microinverters such as the older Enphase M190 can be disastrous for HF, VHF and UHF reception.  As noted in the QST article, the Enphase power optimizers (model number not mentioned) also caused QRM.

Figure 3: The two Tesla Powerwalls, gateway and electrical sub- panels for the system located remotely on the east wall of the house. Click on the image for a larger version

Additionally, it has been observed that the Solaredge inverters - particularly coupled with optimizers - have caused tremendous radio frequency interference:  The aforementioned April, 2016 QST article about solar RFI deals with this very combination.

It probably won't work in all cases.

Compared to some installations that I have seen, my system - or the one in the 2016 article - are very simple cases - and there are a number of practical limitations, which include:

• A "minimum" array size limitation.  Taking the Sunny boy SB5.0 as an example, there is a 90 volt minimum input which means that one would (very conservatively) want at least four 60-cell panels on each circuit.  This limitation may affect what areas on a roof may be candidates for placement of solar panels, reducing the total system capacity as compared to what might be possible with individually-optimized panels.

• Systems with complicated shading.  If there are a number of trees - or even antennas and structures - portions of sub-strings may be shaded, causing reduction in output and compared to individually-optimized panels, series-strings are at a disadvantage, but careful selection of sub-string geometry can help.  For example, if a tower shades a series of panels during the period of highest production, placing all of those panels on one particular string can help isolate the degradation - but this sort of design consideration will require careful analysis of each situation.

Final words:

The design, configuration and layout of a home (or any) PV system is more complicated than depicted here and any system to be considered would have to take into account.  While I am certain that there are other ways to make an "RF Quiet" PV system, this article was intended to be limited to configurations and equipment with which I have direct experience.

Again, the likelihood of finding a "solar professional" who thoroughly understands RFI issues and knows which type of equipment is RF-quiet is unlikely, so it is up to you as the potential recipient of QRM to do the research.

Other articles at this blog on related topics:

• The Solar Saga Part 1: Avoiding Interference - (Why I did not choose Microinverters) - link
  
• The Solar Saga Part 2:  Getting the system online - link
• Does the Tesla Powerwall 2 produce Radio Frequency Interference? - link
• Reducing RFI from the Tesla Powerwall 2 - link 
• Tesla Powerwall susceptibility to RF Transmissions - and how to deal with it - link 
  

This page stolen from ka7oei.blogspot.com

[End]

The SDRPlay RSP2pro (left) and RSP1a receivers (right)

This receiver shares a similar internal architecture of similar devices such as the RTL-SDR dongle and the AirSpy in that an analog frequency converter (mixer) precedes the analog-to-digital converter:  In the case of the SDRPlay, the frequency to which the receiver is tuned is (usually) converted to baseband I/Q signals, with the "center" frequency being at zero Hz (DC). ¹

Note:

For the purposes of this discussion, there is no difference between the RSP1a and some of the other receivers in the product lineup (e.g. RSPDuo, RSPdx and the discontinued RSP1, RSP2 and RSP2pro) in terms of harmonic response across the 2-30 MHz range as they all have about the same 12 MHz and 30 MHz cut-off frequencies on their input filtering - properties that would affect HF reception across the 2-30 MHz range in terms of harmonic response.

This issue was noted at least as far back as 2017 in the SDRPlay forum - See this thread:  https://www.sdrplay.com/community/viewtopic.php?t=2280

Imperfect mixers

By its nature, a frequency mixer is a non-linear device.  Ideally, the two frequencies applied to a mixer would yield just two more - the sum and difference.  For example, if we applied a 5 MHz signal and a 1 MHz signal to a mixer, it would output both the sum of 6 MHz and the difference of 4 MHz - and this is true, but there's more to the story.

In our example - with a real-world mixer, we will also get additional products - including those related to the harmonics of the local oscillator and the applied signal.  Because of this, we will see weaker signals at:

• 11 MHz (2 * 5 MHz + 1 MHz) 
• 9 MHz (2 * 5 MHz - 1 MHz) 
• 7 MHz (5 MHz + 2 * 1 MHz) 
• 3 MHz (5 MHz - 2 * 1 MHz) 
• And so on.

Typically, these "other" signals will be quite a bit weaker than the original - but they will still be present, possibly at a high enough level to cause issues such as spurious signals - a problem with both receivers and transmitters.  Typically, these are tamed by proper design of the mixer, proper selection of frequencies and careful filtering around the mixer to limit the energy of these "extra" signals.

Note:  There will be a response at 5x the center frequency as well, but it is suppressed better than the 3x response by the mixer and - for the 80 meter amateur band and higher - these responses are suppressed reasonably well by the filtering.

SDRPlay's poor harmonic response suppression on 80 meters and below.

ANY receiver will experience spurious responses related to mixing products.  Typically, filtering is employed to remove/minimize such responses, but for a wide-bandwidth receiver such an SDR, doing this is complicated by the fact that being able to cover wide swaths of bandwidth would ideally require a large number of overlapping filters.

An example of a radio where this is done - albeit of different architecture - is the Icom IC-7300 which has nine overlapping band-pass filters that cover 160 through 10 meters.  While the reasons for the '7300 having many filters has as much to do with its being a "direct sampling" ² type of SDR, good filtering on the signal path of any type of receiver - SDR or "HDR" (Hardware Defined Radio - or an "old school" analog type) is always a good idea

If this many filters had been implemented on the SDRPlay, there would be enough filtering to prevent a significant harmonic response.  In the case of the RSP1a, this was not done - partly to allow 5-8 MHz of continuous coverage without being significantly impacted by the filters in many cases, but more likely it was done due to practical reasons of economics ³ :  There are just three filters used for covering all of the "HF" amateur bands 160 through 10 meters:  One that covers up to 2 MHz, another that covers 2-12 MHz and third that covers 12-30 MHz:  This information is covered in the RSP1a technical information document ( https://www.sdrplay.com/wp-content/uploads/2018/01/RSP1A-Technical-Information-R1P1.pdf )

The sensitivity to harmonics was tested with the RSP1a's local oscillator (but not necessarily the virtual receiver) tuned to 3.7 MHz ⁴ .  For reasons likely related to circuit symmetry, it is odd harmonics that will elicit the strongest response which means that it will respond to signals around (3.7 MHz * 3) = 11.1 MHz.  "Because math", this spurious response will be inverted spectrally - which is to say that a signal that is 100 kHz above 11.1 MHz - at 11.2 MHz - will appear 100 kHz below 3.7 MHz at 3.6 MHz.  (It's likely that there are also weaker responses at frequencies around 5 times the local oscillator, but these are - for the most part - adequately suppressed by the filtering.)

In other words, the response to spurious signals follow this formula:

Apparent signal = Center frequency + ((Center frequency * 3) - spurious signal) )

Where:

• Center frequency = The frequency to which the local oscillator on the RSP is tuned.  In the example above, this is 3.7 MHz.
  
• Spurious signal = The frequency of spurious signal which is approximately 3x the center frequency.  In the example above, this is 11.2 MHz.
  
• Apparent signal = Lower frequency where signal shows up.   In the example above, this is 3.6 MHz.
  

While these spurious responses may not be too much of a problem for the casual user, it will be necessary to add additional filtering to allow the RSP1a to function on par with a modern, SDR receiver from one of the major manufacturers.

Unfortunately, the filtering in the RSP1a is not sufficient in the 80 meter case mentioned above as it doesn't have octave filters (or similar) - but what about 60 or 40 meters?

The table above answers this question.  In the case of 60 meters - with the receiver tuned to 5.3 MHz - our 3rd harmonic will land on 15.9 MHz.  Based on measurements of the receiver the response of signals around 15 MHz - which corresponds to the 19 meter Shortwave Broadcast Band - will be a bit more than 40 dB down from 40 meters with about 20 dB of this being due to the roll-off of the 2-12 MHz filter - but because this frequency range is inhabited by very strong shortwave broadcasters they are likely to still be quite audible around 60 meters.

The situation is a bit better for 40 meters where the 3rd harmonic is around the 15 meter band.  There, the 2-12 MHz filter knocks signals down by 50dB or more, putting them about 70dB below the 40 meter response - on par with about any respectable receiver.

What this means is that for amateur bands below 40 meters it is suggested that additional filtering be applied.

The best solution - and recommended for any software-defined radio (or even older "hardware-defined radios") is to have band-pass filter designed for the specific amateur band in question. This will not only significantly attenuate the harmonic response, but it will also reduce the total amount of RF energy entering the receiver, reducing the probability of overload.  The obvious down-side is that it will reduce the flexibility of the receiver in that unless you change/remove it, you won't be able to receive signals well outside the filter's design range.

Another possibility is to add a low-pass filter that is designed to cut off signals above the band of interest.  For example, if you have a filter that cuts off sharply above 8 MHz, you will be able to tune 80-40 meters and get reasonable attenuation of the 3rd harmonic response across this entire frequency range.

In the case of 160 meters the RSP1a will automatically select the 0-2 MHz low-pass filter and the 3rd harmonic response will be a respectable 50-ish dB down, depending on frequency.

On 20 meters - where the 3rd harmonic is around 42 MHz - the "12-30 MHz" filter will be selected, but the published response of this filter shows that at 42 MHz its attenuation will be quite limited.  Practically speaking, it is unlikely that there will be any signals in this frequency range so there being "only" 20-30dB of attenuation is unlikely to cause a problem in most cases, but one should be aware of this.

What can be done:

In short, none of the currently-made SDRPlay receivers - by themselves - will offer very good performance in terms of harmonic rejection between 2 and 5 MHz and it will be particularly bad on the 80 meter band where strong 25 meter SWBC signals can appear:  It is interesting that the ARRL review of the RSPdx (Link here) didn't catch this issue.

It is unfortunate that the designers of the SDRPlay receivers did not add at least one additional low-pass filter in the signal path to quash what is a rather strong response in the 2-6 MHz range - particularly on 80 meters, one of the most popular bands.  A low-pass filter with a cut-off frequency of 6 MHz (with attenuation becoming significant above 7 MHz) would ameliorate the harmonic response when tuning across this band.  This problem is made even worse by the fact that even antennas that aren't particularly resonant at their harmonic responses (e.g. the antenna for 80 meters) will likely do quite a decent job of receiving signals in the 11-12 MHz area.

The only real "fix" for this is to install additional filtering between the SDRPlay receiver and the antenna.  If single-band operation is all that is desired, the best choice will be a band-pass filter designed for the frequency range in question ⁵ - but unless you are dedicating the receiver just for that one band, this isn't really desirable unless you can easily switch/bypass the filter when tuning elsewhere.

A more flexible solution would be to use a low-pass filter.  As we noted above, the 12 MHz roll-off of the built-in (2-12 MHz) filter just doesn't do much to suppress signals from 20 meters, but if we had a filter that had a sharp cut off beginning, say, at 8 MHz, we could use it for 80, 60 and 40 meters - such a filter is depicted schematically, below:

8 MHz low-pass filter schematic, designed using ELSIE

 If one owns a receiver with a waterfall display, the increased cluttering of the 12 and 10 meter bands with weird "swooping" signals could not have gone unnoticed.  Take, for example, this recent snapshot of the lower portion of 10 meters from the waterfall of WebSDR #5 at the Northern Utah WebSDR (Link)

Figure 1: 10 Meters as seen on a beam antenna pointed toward Asia showing QRM from a large number of different sources - presumably dielectric heaters/welders/seamers.  These things radiate badly enough that they should have their own callsigns, right? Click on the image for a larger version.

In looking at this spectral plot - which comes from an antenna oriented to the Northwest (toward Asia and the Pacific) one could be forgiven for presuming that someone had somehow connected a can of "Silly String" to their coax and was squirting noodles into the ionosphere!

What, specifically, are we looking at?

Across the entire spectrum plot one can see these "curved" signals, some of them - like that near the bottom, just above the cursor at 28374 kHz - are quite strong while there are many, many others that are much weaker, cluttering the background.  These signals contrast with normal SSB and CW signals - the former being seen clustered around 28500 and the latter around 28100 kHz - which are more or less straight lines as these represent transmissions with stable frequencies.

What are these from?  The general consensus is that these are from "ISM" (Industrial, Scientific and Medial) devices that nominally operate around 26957 kHz to 27283 kHz.  Clearly, the waterfall plot shows many devices outside this frequency range.

What sort of devices are these?  Typically they are used for RF heating - most often for dielectric sealers of plastic items such as bags, blister packs - but they could also be used in the manufacture of items that require some sort of energetic plasma (e.g. sputtering metal, etching) in any number of industrial processes.

Where are they coming from?

The simple answer is "everywhere" - but in terms of sheer number of devices, it's more likely that much of the clutter on these bands originates in Asia.  Consider the above spectral plot from an antenna located in Utah pointed at Asia - but then consider the plot below, taken at about the same time from an antenna that is pointed east, across the continental U.S. and Canada - WebSDR #4 at the Northern Utah WebSDR (link):

Figure 2: 10 Meters on a beam pointed toward the U.S. Click on the image for a larger version.

To be absolutely fair, this was taken as the 10 meter band was starting to close across the U.S, but it shows the very dramatic difference between the two antenna's directionality, hinting at a geographical locus for many of these signals.

Further proof of the overseas origin of these signals can be seen in the following plot:

Figure 3: Spectrum from AM demodulation of some of the signals of Figure 1 showing 50/100 Hz mains energy. Click on the image for a larger version.

This plot was taken by setting the WebSDR to AM and setting for maximum bandwidth, tuning onto a frequency where several of these "swoops" seen in Figures 1 and 2 are recurring and then, using a virtual audio cable, feeding the result directly into the "Spectran" program (link).

As expected this plot shows a bit of energy at the mains harmonic frequencies of 120, 240 and 360 Hz owing to the fact that this antenna points into slightly-noisy power lines operating at the North American 60 Hz frequency - but on this plot you can also see energy at 50 and 100 Hz, indicative of a lightly-filtered power supply operating from 50 Hz power mains - something that is NOT present anywhere in North America.

Based on other reports (IARU "Intruder Watch", etc.) a lot of these devices seem to be located in Asia - namely China and surrounding countries where one is more likely to experience lax enforcement of spurious radiation of equipment that is manufactured/sold in those locales.

Why the "swoop", "curve" or "fishook" appearance seen in Figure 1?  If these devices were crystal controlled and confined to the nominal 26957 kHz to 27283 kHz ISM frequency range, we probably wouldn't see them in the 10 meter amateur band at all, but many of these devices - likely "built to cost" simply use free-running L/C oscillators that are accurate to within 10-15% or so:  As these oscillators - which are likely integral to the power amplifier itself (perhaps self-excited) - warm up, and as the industrial processes itself proceeds (e.g. plastic melts, material cures, glue dries) the loading on the RF output of this device will certainly change, and this results in an unstable frequency.

Why do they radiate?

Ideally, the RF would be contained to the working area and in the past, reputable manufacturers of such equipment would employ shielding of the equipment and filtering of power and control leads to confine the RF within.  But again, such equipment is often "built to cost" and such filtering and shielding - which is not necessary for the device to merely function is often omitted.

Can we find and fix these?

In this U.S. and parts of Europe such sources are occasionally tracked down and RF interference mitigated - either voluntarily or with "help" from the local regulator - but the simple fact is that the intermittent nature of these sources - and the fact that they radiate on frequencies that are prone to good propagation when the sun is favorable to such - makes them very difficult to localize.  If the signal source is coming from halfway around the world, there's likely nothing that you can do other than point your directional antenna the other way!

If it so-happens that you can hear such a signal at your location at all times of the day - regardless of propagation - you may be in luck:  There may be a device with a short distance (a few miles/km) of your location - and perhaps you can make a visit and help them solve the problem.

* * * * * * * 

Related article:

•  27 MHz ISM Fish Hook Swisher Interference RFI EMI RF HF Sealers Heaters Welders  - link - A thread on this topic found at "HF Underground".

This page stolen from ka7oei.blogspot.com

[End]

Figure 1: Close-in responses of various filter combinations Yellow:  Duplexer-only Magenta:  Bandpass-only Cyan:  Duplexer + Bandpass Click on the image for a larger version.

There is a follow-up article to this one - "A simple VHF notch cavity from scraps of (large) heliax" - link.

The case for bandpass filtering

If you operate a repeater - or even a simplex radio such as a Packet node - that is located at a "busy" radio site, you'll no doubt be aware of the need for cavity-based filtering.

In the case of a repeater, the need is obvious:  Filtering must be sufficiently "strong" to keep the transmit signal out of the receiver, and also to remove any low-level noise produced by the transmitter that might land on the receive frequency.

In the case of a packet or simplex node of some sort, a simple "pass" cavity is often required at a busy site to not only prevent its receiver from being overloaded by off-frequency signals, but also be a "good neighbor" and prevent low-level signals from your transmitter from getting into other users' receivers - not to mention the preventing of those "other" signal from getting back into your transmitter to generate spurious signals in its own right.

Comments:

• In this discussion, a "band pass" filter refers to the passing of ONLY a narrow range of frequency near those of interest and at odd multiples of the lowest resonant frequency - but nothing else.

• It is HIGHLY RECOMMENDED that anyone attempting to construct this type of filter get and learn to use a NanoVNA:  Even the cheapest units (approximately $50US) - when properly set up - will be capable of the sorts of measurements depicted in this article.

A Band-Pass/Band-Reject (BpBr) duplexer may not be what you think!

A common misconception is that a typical repeater duplexer - even though it may have the words "band pass" written on its label or in its specifications - has a true "band pass" response.

The problem becomes more apparent when we look over a broader frequency range.  Figure 2 shows the same hardware, but over a span of about 30 MHz to 1 GHz.

Figure 2: The same as in Figure 1 except over a wider frequency range showing the lack of off- frequency rejection of a "BpBr" duplexer (Yellow) that is significantly mitigated by the addition of a band-pass filter (Cyan) Click on the image for a larger version.

Keeping an eye on the yellow trace, you'll note that over most of the frequency range there is very little attenuation.  What this means is that the "BpBr" filter doesn't exhibit a true pass response once you get more than a few MHz away from the design frequency.

I've actually had arguments with long-time repeater owners that disagreed with this assertion, but hadn't actually "swept" a duplexer over a wide frequency range:  These days, with the availability of inexpensive test equipment like the NanoVNA, there's no good excuse for not determining this for yourself!

For more about this, see the related article linked here.

Why is this a problem?

In the "old days" radios that you would use at a repeater site were typically cast-off mobile radios - and even if you had a repeater, it was typically based on a mobile design.  These older radios - often from the 80s or earlier - typically used a bank of narrowband  (often Helical) filter elements, each tuned to the frequency of interest:  If several frequencies were used, the system planners often placed then near each other so that they could be covered by the receivers' narrow filters without undue attenuation and because of this, one could "get away with" a duplexer with filtering that didn't offer a "true" pass-band response.

Most modern radios used in amateur repeaters are "broadband" in nature meaning that they often have rather wide receiver front-end filters:  It is not practical to have electronically-tuned filters that are anywhere near as narrow as the Helical filters of the past which means that they simply lack the filtering to reject strong, off-frequency signals.

The poor filtering of some "new" radios:

When a modern radio is dropped in place of an old radio at a "busy" site with lots of other transmitters, disappointment is sometimes the result:  The "new" radio may seem less sensitive than the old one - or it might seem that sensitivity varies over time.  In reality, the "new" radio may well be being overloaded by the off-frequency signals that the old radio's resonator-based front-end easily filtered.  What's worse is that the precise nature of this overload condition may be masked by the use of subaudible tones or digital tone squelch - and if this is a digital radio system like D-Star, Fusion or DMR, there may be no obvious clues at all as to the problem at hand unless one has the ability to measure and monitor the analog "baseband" from the receiver itself.

To be sure, if the receiver in question can operate in carrier-squelch analog mode, the usual techniques to determine overload (Iso-Tee measurements, injection of a weak carrier and observing SNR, etc.) may be employed to determine if there is an issue - but this, too, may be misleading as problems may be intermittent, showing up only when a combination of transmitters key up.

A simple pass cavity:

While not a panacea, the use of a simple pass-only cavity can go a long way to diagnose - even solve - some chronic overload issues - particularly if these have arisen when old gear was replaced.  Suitable pass cavities are readily available for purchase new from a number of suppliers and used from auction sites - they are also pretty easy to make from copper and aluminum tubing - if you have the tools.  Because of the rather broad nature of a typical pass cavity, temperature stability is usually not much of an issue in that its peak could drift hundreds of kHz and only affect the desired signal by a fraction of a dB.

Another material that could be used to make reasonable-performance pass cavities is larger-diameter hardline or "Heliax" ^(tm).  Ideally, something on the order of 1-5/8" or larger would be used owing to its relative stiffness and unloaded "Q" and either air or foam dielectric cable may be used, the main difference being that the "Q" of the foam cable will be slightly lower and the cavity itself will be somewhat shorter.

Figure 3: Cutting the (air core) cable to length Click on the image for a larger version.

The "Heliax bandpass cavity" described here can be built with simple hand tools, and it uses a NanoVNA for tuning and final adjustment.   While its performance will not be as good as a larger cavity, it will - in many cases - be enough to attenuate strong, out-of-band signals that can degrade receiver performance.

Using 1-5/8" "Heliax":

The "cavity" described uses 1-5/8" air-core "Heliax" - and it is necessary for the inner conductor to be hollow to accommodate the coupling capacitors.  Most - but not all - cable of this size and larger has a hollow center conductor.  Cables larger diameter than 1-5/8" should work fine - and are preferred - but smaller than this may not be practical - both for reasons of unloaded "Q" and also if the center conductor is solid or if its inside diameter cannot accommodate the coupling capacitors described later on.

Note:  The KF6YB article linked below contains some length calculators for coaxial cable.

Preparing the "shorted" end:

For 2 meters, a piece of cable 18" long was cut.  For the air dielectric, it's recommended that one cuts it gently with a hand saw rather than a power tool as the latter can "snag" and damage the center conductor.

Figure 4: The "shorted" end of the stub with the slits bent to the middle and soldered to the center conductor. Click on the image for a larger version.

For the "cold" (e.g. shorted) end, carefully (using leather gloves) remove about 3/4" (19mm) of the outer jacket and then clean the exposed copper shield with a wire brush, abrasive pad and/or sand paper.  With this done, use a pair of tin snips cut slots about 1/2" (12mm) deep and 1/4" (6mm) wide around the perimeter.  Once this is done, use a pair of needle nose pliers and remove every other tab, resulting is a "castellated" series of slots.  At this point, using a pair of diagonal pliers or a knife, cut away some of the inner plastic dielectric so that it is about 1/2" (12mm) away from the end of the center conductor.

Now, clean the center conductor so that it is nice and shiny and then bend the tabs that were cut inwards so that they touch the center conductor.  Using a powerful soldering iron (I used a 150 watter) or soldering gun - and, perhaps a bit of flux - solder the shield tabs to the center conductor all of the way around.  It's best to do this with the section of coax laying on its side so that hot solder/metal pieces do not end up inside the coax - particularly if air-core cable is used.  If you used acid-core flux, carefully remove it before proceeding.

With one end of the cable shorted you can trim back any protruding center conductor and file any sharp edges - again taking care to avoid getting bits of metal inside the cable or embedded in the foam.  At some point, you should cover the shorted end with RTV (silicone) and/or good-quality electrical tape to prevent contamination by dust or insects.

Preparing the "business" end:

Figure 5: The "coupling tubes" soldered in place which receive the wires for coupling in/out. Click on the image for a larger version.

Making coupling capacitors:

We now need to make two capacitors to couple the energy from the "in" and "out" connectors to the center resonator and for this, I cut two 3" (75mm) long pieces of RG-6 foam TV coaxial cable and from each of these pieces, I removed and kept the center conductor and dielectric - removing any foil shield and then stripping about 1/2" (12mm) of foam from one end of each piece.

At this point, you'll need some small copper tubing:  I used some 1/4" O.D. soft-drawn "refrigerator" tubing, cutting two 2" (50mm) lengths and carefully straightening them out.  To cut this, I used a rotary pipe cutting tool which slightly swedged the ends - but this worked to advantage:  As necessary, I opened up the end cut with the deburring blade of the rotary cutting tool just enough that it allowed the inner dielectric of the RG-6 to slide in and out with a bit of friction to hold it in place..

Comment:

The use of 1/4" (6mm) O.D. copper pipe and RG-6 center conductor/dielectric isn't terribly critical:  A different-sized copper or brass pipe could be used as long as two parallel pieces will fit inside the center conductor of the Heliax - and that the chosen center conductor and dielectric of the coax you use to make the capacitor will fit somewhat snugly inside it.

The reason for the two copper tubes is to prevent the two capacitors made from the center of the RG-6 from coupling directly to each other as all energy must first resonate the center conductor.  Using these tubes - soldered to the center conductor/resonator - prevents such direct coupling, and it offers good mechanical stability.

Figure 6: The PC Board plate soldered to the end of the coax. Click on the image for a larger version.

Using a hot soldering iron or gun, solder the two straightened pieces of tubing together, in parallel, making sure that the ends of the tubing that you adjust to snugly fit the outside diameter of the piece of RG-6 are at the same end.  Once this is done, insert the two parallel pieces of tubing inside the Heliax's center conductor and solder them, the ends flush with the end of the center conductor, taking care not to heat them enough that they unsolder from each other:  A pair of sharp needle-nose pliers to hold them in place is helpful in this task.

Making a box:

On the "business" (non-shorted) end of the piece of cable we need to make a simple box with a solid electrical connection to the outer shield to which we can mount the RF connectors with good mechanical stability.  For the 1-5/8" cable, I cut a piece of 0.062" (1.58mm) thick double sided glass-epoxy circuit board material into a square that was 3" (75mm) square and using a ruler, drew lines on it from the opposite corners to form an "X" to find the center.

Using a drill press, I used a 1-3/4" (45mm) hole saw to cut a hole in the middle of this piece of circuit board material, using a sharp utility knife to de-burr the edges and to enlarge it slightly so that it would snugly fit over the outside of the cable shield:  You will want to carefully pick the size of hole saw to fit the cable that you use - and it's best that it be slightly undersized and enlarged with a blade or file than oversized and loose.

Figure 7: Bottom side of the solder plate showing the connection to the coax. Click on the image for a larger version.

After cleaning the outside of the coaxial cable and both sides of the circuit board material, solder it to the (non-shorted) end on both sides of the board, almost flush with just enough of the shield protruding through the top to solder it.  For this, a bit of flux is recommended, using a high-power soldering iron or gun - and it's suggested that it first be "tacked" into place with small solder joints to make sure that it is positioned properly.

When positioning the box, rotate it such that the two "capacitor tubes" that were soldered into the center conductor are parallel with one of the sides of the square - this to allow symmetry to the connectors:  This is depicted in Figure 8 where the left-hand and right-hand tubes (more or less) line up with their respective coaxial connectors.

Adding sides and connectors:

With the base of the box in place, cut four sides, each being 1-3/8" (40mm) wide and two of them being 3" (75mm) long and the other two being 2-1/2" (64mm) long.  First, solder the two long pieces to the top, using the shorter pieces inside to space and center them - and then solder the shorter pieces, forming a five-sided (base plus four sides) box atop the piece of cable. As seen in the photo, the "short" sides are parallel to the two tubes in the center conductor.

As can be seen in Figure 8, there is a piece of PC board material about 1/4-3/8" (6-10mm) wide that goes between the two walls with the BNC connectors.  This piece provides a bit of stiffening, minimizing flexure of the two walls with the connectors when cables are connected which could change the orientation of the two "RG-6 capacitors" - and it provides a small degree of shielding between the input and output wires that form these capacitors.

Figure 8: Inside the box with coupling/tuning stubs and lines and stiffening bar installed.  Note the orientation of the tubes. Click on the image for a larger version.

As can be seen in the picture, BNC connectors were used as they were convenient, but "N" type, SMA or even UHF connectors could be used - but the use of BNC connectors will be described.

The BNC connectors were mounted on opposite sides of the box, approximately 3/8" to the left of the center line and 3/4" from the bottom.  As can be seen in the photo, the connectors were mounted in the "short" wall of the box such that our "RG-6" capacitors more or less line up with the capacitor tubes.

Now, insert the ends of the RG-6 center conductor into the "capacitor tubes" and, bending the top in an "L" shape, solder the end with the exposed center conductor to the coaxial connectors. 

Also visible in Figure 8 - just to the right of the center conductor - are two pieces of copper strip, each about 3/4" (20mm) wide - one is soldered to the center conductor and the other to the ground inside the box.  These two tabs form a very simple capacitor which may be used to "fine tune" the center frequency of the pass response by bending them nearer/closer to each other.  While most of the adjustment of the center tuning will occur as one slides the two capacitors (made from RG-6) in and out, this method may also be used to provide a bit of additional tuning range.

Preliminary adjustment:

It is best to first set the RG-6 capacitors to obtain the desired pass response, taking into account the desired band-pass frequency, but once one has done this - and if the pass frequency is too high - one would then add the "copper tab" capacitors - perhaps more than one set.  If the frequency is too low, see if you can obtain a suitable passband width (and acceptable insertion loss) by pulling these coupling capacitors out to raise the frequency:  For most applications, an insertion loss of even 1 dB will not appreciably reduce receiver performance - particularly if the local noise at the site is rather high from other users and especially if the use of a band-pass filter like this is intended to minimize desense, anyway.

It is best to make them from metal (copper, brass) that is thick enough to not be springy on their own:  Saving a piece and flattening the copper material from the shield of the 1-5/8" coax as you prepare it for use is suggested.

At this point we are ready to do some preliminary tuning - and this will require a NanoVNA or similar:  It is presumed that the builder will have familiarity with the NanoVNA to make S11 VSWR and S12 insertion loss measurements on an instrument that has been properly calibrated at the frequency range in question.

Setting the NanoVNA to measure both VSWR and through-loss over a span of 130-160 MHz, connect it to the cable and you should see a pass response somewhere in the frequency range and if all goes well, the peak in the pass response will be somewhere in the 130-140 MHz range.

Adjusting the center frequency and passband response is an iterative process as reducing the coupling by pulling out the capacitors (the RG-6 center conductor) will also increase the frequency.  Practically speaking, only about 3/8"-1/2" (9-12mm) of center conductor is needed at most to attain optimal coupling so don't be afraid to pull out more and more of the capacitors.

A bit of experimentation is suggested here to get the "feel" of the adjustment - and here are a few pointers:

• Figure 9: The band-pass filter sitting against a Sinclair Q2220E 2-meter Duplexer - a good combination for receiver protection at a busy repeater site! Click on the image for a larger version.

  Lowest SWR is obtained with the coupling capacitors are identically adjusted.  If the SWR isn't below 1.5:1, try pulling out or pushing in one of the capacitors slightly to determine the effect - but move it only about 1/16" in each iteration.  Generally speaking, pulling one out slightly is the same as pushing the other in slightly in terms of reducing VSWR.
• The passband response will be narrower the less of the RG-6 center conductor is in the capacitor tubes - but the insertion loss will also go up.
  
• The frequency will go up the more the passband response is narrowed by reducing the coupling capacitors.
• With the lengths given (e.g. 17" for air-core 16-1/8" for foam core) the passband will be within the 2 meter band with the amount of coupling that will yield about 0.5-0.6 dB insertion loss.  To a degree, you can "tune" the center frequency of the cavity by adjusting the coupling.  It is recommended that you first tune for the bandpass response - and then tune it to frequency: See below for additional comments.
  
• It is recommended that you use a little coupling as needed to obtain the desired response.  For example, if the cavity is "over coupled", the insertion loss will be about 0.5dB, but this will go up only very slightly as the coupling is reduced and the response is narrowed.  At some point the insertion loss will start to go up as the passband is further-narrowed.
• As the coupling capacitor is pushed in, the resonant frequency will go down.  If, even with the "tuning capacitor" (the copper strips) are minimized in coupling,  the frequency is too low, try pulling the RG-6 capacitors out slightly to move it up in frequency:  It's easy to accidentally "over couple" the cavity by pushing them in too far and causing it to tune low.   It's likely that you can pull more of the RG-6 capacitors out and reducing coupling than you might first think and still have acceptably-low insertion loss - and doing so will narrow the passband response and improve ultimate off-frequency attenuation.
  

As mentioned above, it's recommended that the approximate passband width be set with the capacitors and if all goes well, the pass frequency can be adjusted with just the adjustment to the coupling with only a slight change in overall bandwidth.  If, however, the desired "narrowness" results in a pass frequency above that which is desired, a simple "tab" capacitor can be constructed as shown in the photo.

Figure 10: The "close-in" response of the band-pass cavity. With the current settings providing a bit less than 0.5dB of attenuation at the center, it's rejection at the edges of the U.S. 2 meter band (144-148 MHz) is a bit over 8 dB. Click on the image for a larger version.

With the preliminary tuning done, a bit of reinforcement of the box is suggested:  A strip of copper circuit board material 3/8"-1/2" (9-12nn) wide is soldered between the inside walls of the box with the RF connectors.  This strip minimizes the flexing of the walls with the RF connectors due to stresses on the connected cables which can change the orientation of the coupling capacitors and cause slight detuning.

With this reinforcement in place, do a final tweaking of the bandpass filter's tuning.

Final assembly:

As noted earlier, it's strongly suggested that the shorted end of the cavity be covered to prevent debris and insects from entering either the center conductor or, especially, the space between the shield and center conductor.  This may be done using electrical tape or RTV (Silicone) adhesive.

Figure 11: A wider sweep showing the rejection at and below the FM broadcast band and up through 225 MHz.  This filter, by itself, provides over 40 dB rejection at 108 MHz Click on the image for a larger version.

Additional comments:

While the performance will vary depending on the coupling and tuning, the prototype, tuned for a pass response at 146.0 MHz, performed as follows:

• Insertion loss at resonance:  <0.5dB
• -3dB points:  -88 kHz and +92 kHz
  
• -10dB points:  -2.5 MHz and +2.75 MHz
• -20dB points:  -7.6 MHz and +11 MHz
• 2:1 VSWR bandwidth:  600 kHz
• Loss <=108 MHz:  40dB or greater

More detail about the response of this cavity filter may be seen in figures 10 and 11.  In the upper-left corner of each figure may be found the measured loss and VSWR at each of the on-screen markers.

If a higher insertion loss can be tolerated, the measured bandwidths will be narrower.  Depending on the situation, an extra dB or two of path loss may be a reasonable trade-off for improved off-frequency rejection - particularly on a noisy site where the extra loss won't result in a degradation of system sensitivity due to the elevated noise floor and the improved selectivity reduces off-frequency signals even more.

As with any cavity-type filter, there is a bit of fragility in terms of frequency stability with handling.  If, after it is tuned this - or any cavity filter - is dropped or jarred strongly, the tuning should be re-checked and adjusted as necessary.

There's no reason why this cavity couldn't be used for transmitting, although using the materials described (e.g. the center conductors of RG-6) I would limit the power to 10-15 watts without additional testing.

As it is, this band-pass filter - in conjunction with a conventional 2-meter duplexer - can provide a significant reduction in off-frequency energy that could degrade receiver performance.  As can be seen in Figure 2, the pass cavity may still pass energy from odd-order (3rd, 5th) harmonics that may fall within commercial/70cm and TV broadcast frequencies - but the addition of a VHF low-pass filter - perhaps even the VHF side of a VHF/UHF mobile diplexer - would eliminate these responses.

To be a good neighbor on a busy site it's strongly recommended that a pass cavity also be installed on the transmit side, along with a ferrite isolator (e.g. circulator with dummy load) to deal with signals that may enter into the transmitter's output stage and mix, causing intermodulation distortion and interference - both to your own receiver and those of others. 

* * * * * * 

Specific use cases:

Reduction of ingress from FM broadcast transmitters:

The most obvious use case would be the filtering of co-located FM broadcast transmitters.  Despite being about 50 MHz off-frequency, a nearby FM transmitter - which often runs hundreds if not thousands of watts - can couple into a 2 meter antenna with sufficient energy to overload a receiver, causing the appearance of distorted audio from the wideband FM modulation of the broadcast transmitter to appear on the receiver.  I have been on sites where the measured power at the receiver terminals, after passing through the 2 meter duplexer, have been on the order of 10 to 20 dBm (10-100 milliwatts) and actually registered as reflected power on an SWR bridge.

As can be seen from the above graphs and measurement, the filter described is capable of reducing this signal by at least 40 dB - likely enough to prevent gross overload of the receiver in question.

Reduction of commercial high-band VHF signals:

While less common these days, there are still systems like community repeaters operating above the 2 meter band in the 150-170 MHz range.  These, too, can cause receiver degradation and the fact that these signals may be intermittent (e.g. a repeater that isn't used too often) can often frustrate the analysis of intermittent "desense" issues.  Even this humble cavity is capable of reducing such a signal by about 20 dB at and above 155 MHz - more, if one were to reduce the coupling (and response bandwidth) of the cavity:  In severe cases, the slightly higher insertion loss of the filter (say, 1.5 dB instead of less than 0.5 dB) may well be worth the trade-off in off-frequency rejection.

* * * * * *

"I have 'xxx' type of cable - will it work?"

The dimensions given in this article are approximate, but should be "close-ish" for most types of air and foam dielectric cable.  While I have not constructed a band-pass filter with much smaller cable like 1/2" or 3/4", it should work - but one should expect somewhat lower performance (e.g. not-as-narrow band-pass with higher losses) - but it may still be useful.

Because of the wide availability of tools like the NanoVNA, constructing this sort of device is made much easier and allows one to characterize both its insertion loss and response as well as experimentally determining what is required to use whatever large-ish coaxial cable that you might have on-hand.

"Will this work on (some other band)?"

Yes, it should:  Notch-only filters of this type were constructed for a 6 meter repeater - and depending on your motivation, one could also build such things for 10 meters or even the HF bands!

It is likely that, with due care, that one could use these same techniques on the 222 MHz and 70cm bands provided that one keeps in mind their practical limitations.

* * * * * *

Related articles:

• There is a follow-up article to this one - "A simple VHF notch cavity from scraps of (large) heliax" - link.
  

• Second Generation Six-Meter Heliax Duplexer by KF6YB - link  - This article describes a notch type duplexer rather than pass cavities, but the concerns and construction techniques are similar.
  

• When Band-Pass/Band-Reject (Bp/Br) Duplexers really aren't bandpass - link - This is a longer, more in-depth discussion about the issues with such devices and why pass cavities should be important components in any repeater system.

This article was stolen from ka7oei.blogspot.com

[END]

Note:  As of 9 February, 2022, this signal is still there, doing what it was doing when this post was originally written.

* * *

Listening on 20 meters, as I sometimes to, I occasionally noticed a loud "click" that seemed to pervade the upper portion of the band.  Initially dismissing it as static or some sort of nearby electrical discharge, my attention was brought to it again when I also noticed it while listening on the Northern Utah WebSDR - and then, other WebSDRs and KiwiSDRs across the Western U.S.  Setting a wide waterfall, I determined that the source of this occasional noise was not too far above the 20 meter band, occasionally being wide/strong enough to be heard near the top of the 20 meter band itself.

Figure 1: The carrier in question - with a few "clicks".  In this case, the signal in question was at 14.390 MHz. Click on the image for a larger version.

From central Utah, a remote station with a beam indicates that the bearing at which this carrier peaks is somewhere around northeast to east-northeast, but it's hard to tell for certain because of the normal QSB (fading) and the fact that the antenna's beamwidth is, as are almost all HF beams, 10s of degrees wide.  Attempts were made to use the KiwiSDR "ARDF" system, but because it is effectively unmodulated, the results were inconclusive.

What is it?

The frequency of this signal appears to vary, but it has been spotted on 14.378 and 14.390 kHz (other frequencies noted - see the end of this article) - although your mileage may vary.  If you listen to this signal sounds perfectly stable at any given instant - with the occasional loud "click" that results in what looks like a "splat" of noise across the waterfall display (see Figure 1), with it at the epicenter

Comment:   If you go looking for this signal, remember that it will be mostly unmodulated - and that it will be subject to the vagaries of HF propagation. 

When a weird signal appears in/near the amateur bands - particularly 20 meters - the first inclination is to presume that it is an "HFT" transmitter - that is, "High Frequency Trading", a name that refers not to the fact that they are on the HF bands, but that it's a signal that conveys market trades over a medium (the ionosphere) that has less latency/delay than conventional data circuits, taking advantage of this fact to eke margins out of certain types of financial transactions.  Typically, the signals conveying this information appear to be rather conventional digital signals with obvious modulation - but this particular signal does not fit that profile.  Why blame HFT?  Such signals have, in the past, encroached in the 20 meter band and distrupted communications - see the previous blog post "Intruder at the top of the 20 meter amateur band?" - link.

Why might someone transmit a (mostly) unmodulated carrier?  The first thing that comes to mind would be to monitor propagation:  The amplitude and phase of a test carrier could tell something about the path being taken, but an unmodulated signal isn't terribly useful in determining the actual path length as there is nothing about it that would allow correlation between when it was transmitted, and when it was received.

Except, that this signal isn't unmodulated:  It has those very wideband "clicks" could help toward providing a reference to make such a measurement.

What else could it be?  A few random thoughts:

• Something being tested.  It could be a facility testing some sort of HF link - but if so, why the frequency change from day to day?  The "clicks"?  Perhaps some sort of transmitter/antenna malfunction (e.g. arcing)?
  

• Trigger for high-frequency trading (HFT).  Many high-frequency trading type signals are fairly wide (10 kHz or so) - possibly being some sort of OFDM - but any sort of coding imposes serialization delays which can negate some of the minimization of propagation delay being attained via the use of HF as compared to other means of conveying data over long distances.  Likely far-fetched, but perhaps the "clicks" represent some sort of trigger for a transaction, perhaps arranged beforehand by more "conventional" means.  After all, what possible means of conveying a trigger that "something should happen" exists than a wide-bandwidth "click" over HF?  Again, unlikely - but seemingly so did something like HFT in the first place!  Additionally, it would seem that the "other" HFT signals that had been present have mostly disappeared - to be replaced by, what?  I suspect that they haven't just gone away!
  

A bit of analysis:

A bit of audio of this carrier, complete with "clicks" was recorded via a KiwiSDR.  To do this, the AGC and audio compression were disabled, the receiver set to "I/Q" mode and tuned 1 kHz below the carrier and the bandwidth set to maximum (+/- 6 kHz) and the gain manually set to be 25 dB or so below where the AGC would have been.  Doing this assures that we capture a reference level from the signal itself (the 1 kHz tone from the carrier) at a low enough level to allow for a very much stronger burst of energy (the "click") to be detected without worrying too much about clipping of the receive signal path.

The result of this is the audio file (12 kHz stereo .WAV) that you may download from HERE.

Importing this file into Audacity, we can zoom in on the waveform and at time index 13.340, we can see this:

Figure 2: Zoomed-in view of the waveform from the off-air recording linked above. These "clicks" seem to come in pairs, approximately 1 msec apart, and have an apparent amplitude hundreds of times higher than the carrier itself. Click on the image for a larger version.

Near the baseline (amplitude zero) we see the 1 kHz tone at a level of approximately 0.03 (full-scale being normalized to 1.0) but we can see the "clicks" represented by large single-sample incidents, one of which is at about 0.83.  Ignoring the fact that the true amplitude and rise-time of this "click" is likely to be higher than indicated owing to band-pass filtering and the limited sample rate, we see that the ratio between the peak of the "click" and the sine wave is a factor of 27.7:1 or, converted to a power relationship, almost 29dB higher than the CW carrier.

This method of measuring the peak power is not likely to be very accurate, but it is, if anything, under-representing the amplitude of the peak power of this signal.  It's interesting to note that these clicks seem to come in pairs, separated by 12-13 samples (approximately 1 millisecond - about the distance that it takes a radio signal 300 km/186 miles) - and this "double pulse" has been observed over several days.  This double pulse might possibly an echo (ionospheric, ground reflection), but it seems to be too consistent.  Perhaps - related to the theoretical possibility of this being some sort of HFT transmission - it may be a means of validation/identification that this pulse is not just some random, ionospheric event.

Listening to it yourself:

Again, if you wish to listen for it, remember that it is an unmodulated CW carrier (except for the "clicks") and that you should turn all noise blanking OFF.  Using an SSB filter, these clicks are so fast that they may be difficult to hear, particularly if the signal is weak.  So far, it has been spotted on 14.378 and 14.390 MHz (try both frequencies) which means that in USB, you should tune 1 kHz lower than this (e.g. 14.377 and 14.389) hear a 1 kHz tone.  Once you have spotted this signal, switching to AM may make hearing the occasional "click" easier. 

Remember that depending on propagation, your location - and your local noise floor - you might not be able to hear this signal at all.  Keep in mind that the HF bands are pretty busy, and there are other signals near these two frequencies with other types of signals (data, RTTY, etc.) - but the one in question seems to be an (almost!) unmodulated carrier.

It's likely that this carrier really isn't several hundred kHz wide, so it may not actually be getting into the top of 20 meters, but the peak-to-average power is so high that it may be audible on software-defined radios:  Because the total signal power across 20 meters may be quite low, the "front end AGC" may increase the RF signal level to the A/D converter and when the "click" from this transmitter occurs, it may cause a brief episode of clipping, disrupting the entire passband.

* * * * *

If anyone has any ideas as to what this might be, I'd be interested in them.  If you have heard this signal and have other observations - particularly if you can obtain a beam heading for this signal, please report them as well in the comments section, below.

Updates:

• November, 2022:   As a follow-up, it would seem that the nature of this "clicky carrier" has changed very slightly.  It appears as though the bandwidth of the "click" is now better-contained and is only a few 10s of kHz wide rather than around 100 kHz wide.

  It also appears that other frequencies are being use - including 14.372 MHz.   More frequencies may be used routinely, but I don't monitor this signal frequently.

• December, 2022:  This type of signal was noted on 14.380 MHz - and possibly 14.413 MHz simultaneously, making for a total of at least four frequencies where this type of signal has been observed.

A few weeks ago I was on vacation in remote Eastern Utah - in Canyonlands National Park, to be precise and because we had some "down time" in the evenings, after hiking, after sunset, I was able to set up a portable HF station.  Using the homebrew end-fed halfwave antenna (EFHW) of Mike, K7DOU - one end of the rope tied around a rock laying on a shelf of slick rock some 40 feet above ground level and the other end tied to a bamboo pole attached to my Jeep - I connected my FT-100 through a manual tuner as the VSWR of the EFHW wasn't necessarily very low on some of the higher bands.

Figure 1: 150 Watt Samlex sine wave inverter, sitting on the workbench. Click on the image for a larger version.

For whatever reason, I had brought along my old lap top and sound-card interface so I could work some digital modes, specifically FT-8 - a mode that I was familiar with, but had personally never worked.  The battery in my laptop had discharged, so I needed an alternate source of power and I connected my 150 watt Samlex Sine Wave inverter (a PST-15S-12A) to the battery to power the computer's power supply.

The (expected!) result of this was a tremendous "hash" all across the HF spectrum - an obvious result of the various high-power converters contained within the inverter.  On some bands the interference wasn't too bad, but on others the result was unusable.  While the battery charged, I operated on the band (20 meters, IIRC) that wasn't as badly affected.

I left the inverter running and the laptop battery charging during the cooking and eating of dinner, and with a reasonable amount of power banked I could turn off the inverter and get a zero noise floor while operating.

Why so noisy?

Modern AC inverters first convert the DC input power to something around the peak voltage found on the AC output - typically around 155 volts for 120 volt mains.  This conversion is done using a switch-mode inverter with a transformer, typically operating in the 20-60 kHz range and this output is rather rich in harmonics.

For the less-expensive "modified sine wave" inverters, the DC output is chopped, typically using an "H" bridge switch using FETs (Field Effect Transistors) with the duty cycle being varied to provide the equivalent of a 120 volt sine wave - and this switching can also add a bit of extra RFI, most notably in the form of a "buzz" - but this action produces less energy at radio frequencies than the initial voltage conversion.

The "Sine Wave" inverters perform the same step of producing the high DC voltage, but will chop the output into much smaller bits.  The method that this is done can vary, but it's sometimes done by using a "buck" type switching converter to transform the higher voltage into a varying - usually lower - voltage to simulate a sine wave on the output.  This second conversion adds yet another source of RF interference atop what is likely already the significant source that already present in the high voltage converter.

Comment:  The power converter (wall wart) that I was using to charge my laptop is particularly quiet, so I did verify that the vast majority of noise was, in fact, from the AC inverter.

Figure 2: Various mains filtering components:  All of these are bifilar, common-mode chokes, except for that in the upper-left with is a combination filter and IEC power connector. Click on the image for a larger version.

Quieting the inverter:

Fortunately, the internal space of this inverter wasn't terribly cramped so there was just enough room to add the necessary components to suppress the RF "hash" that was being conveyed on both the DC and AC lines.  While the methods of doing this sort of RF quieting have been discussed in previous blog posts (see the references at the end of this article) I'll review them in detail here.

Snap-on chokes won't do!

It's worth noting (several times!) that simply winding the power cord (DC and/or AC) around a ferrite device (e.g. a clamp-on or even a large toroid) would likely NOT be enough to solve this problem.  While doing so may knock down RFI by, perhaps, 6-10 dB - maybe 20 dB if one is really lucky - this sort of noise egress must often be attenuated by several 10s of dB to effectively quash it.  In other words, knocking down the "grunge" by 1-2 S-units is nice enough, but there will still be a lot of hash left over to bury the weakest signals! 

Internally, this inverter did pass through some rather large ferrite cylinders the DC input and (separately) AC output connections, but this very small amount of inductance would have practically no effect at all at HF - likely having been added to make a dent in the noise at VHF so that it would pass muster when subjected to EMC compliance tests.

Filtering the AC output:

I presumed (but didn't actually measure) that the majority of the noise being radiated would be from the AC output as it is "closest" to the circuits most likely to generate a lot of noise, so I concentrated most of my effort there.

The most helpful component in filtering the mains voltage output is the bifilar choke - several varieties of these being displayed in Figure 2.  This component consists of two windings in parallel on the same ferrite core - typically both leads of the mains voltage.  For the low-frequency AC currents, the halves of the choke carry equal and opposite current so there is no DC component to magnetize the core and reduce its efficacy due to saturation, but because RF energy is likely not flowing in a differential manner as is the AC mains voltage, the inductance of the two parallel windings come into effect - the magnitude of this typically being in the 10s of microHenries to milliHenries range.

Where does one get these things?  They can be found at surplus outlets if you look around, but perhaps the easiest source is from defunct PC power supplies:  These devices, found in supplies made by reputable manufacturers, are typically the first things through which the AC mains voltage pass (after any fusing) before going to the rest of the circuitry.

Figure 3: Schematic of the output filter.  While it's likely that just one bifilar inductor would have sufficed, I decided that since there was room to do so, a second one would be added for even more filtering of the "grunge" that can emanate from such a noisy circuit. Click on the image for a larger version.

This much inductance has significant impedance to RF energy - but inductance alone will have only limited efficacy and intrinsic capacitance of the windings will also reduce the amount of attenuation that would otherwise happen - as would have winding the mains cord/cable on a ferrite toroidal core as noted previously - so capacitors are also required to be placed strategically to help shunt away some of the residue.

Figure 4: The AC output filter in the process of being installed.  L1 and C1-C4 are mounted to the outlet itself while the connection to L2 is made using the orange leads. Click on the image for a larger version.

The diagram in Figure 3 shows the as-installed filter.  As can be seen, two separate bifilar filters (both of them being the sort as seen as the second from the lower-right in Figure 2) were used to maximize attenuation.  In this circuit, C3 and C4 are used to force any RF on the two wires to be common-mode to maximize the efficacy of the bifilar chokes' attenuation and any residual RF - which will be at rather low level and high impedance - will then be shunted to the metal case of the inverter by capacitors C1 and C2.

Figure 4 shows the installation of the filtering components in the inverter.  C1 and C2 are the disk-shaped blue capacitors seen in the upper-left, mounted directly to the inverter's single AC outlet and capacitor C3 is just in "front" of the two round disks, also mounted directly to the socket.  The first inductor, L1, can be seen in the shadows, connected to the outlet with very short, flexible leads to the plug.

Earlier, I had removed this outlet from the body of the inverter and mounted C1, C2, C3 and L1 to it and with a bit of "tetris" action, was able to reinstall the outlet back in place with the components attached.  From that point I installed C4 (to the "other" side of L1) and the (orange) connecting wires from C4 to L2, which is shown floating in space.

You might ask why there isn't another capacitor (like C4) across the "inverter" side of L2 - or other capacitors to ground other than C1/C2:  There is already a degree of filtering on the AC output of the inverter, so there is little point in adding another capacitor like C4.  As for other capacitors to "ground" like C1/C2 elsewhere in the circuitry:  These were deemed unnecessary - and doing so, particularly at the "inverter" side of L4 would simply put relatively strong RF currents onto the ground lead (e.g. inverter's case) - and our cause won't be helped in making RF currents appear where don't need them to be.  

Figure 5: Noise filter on the DC input.  It looks suspiciously like the filter on the AC output - because it's the same type, although the current-carrying capacity of L1 is much higher and the values of the capacitors are orders of magnitude larger. Click on the image for a larger version.

Filtering the DC input:

While I would presume that most of the noise would be emitted via the AC output port, filtering the DC port must be considered as well.  With the inverter's rating being 150 watts, the maximum current on the AC output would be around 1.25 amps and rather light-gauge wire could be used in the inductors - but because this same power level represents 12.5 amps at 12 volts (likely more if the battery voltage is on the low side) the filtering inductance must be made using much larger wire.

Rummaging around in my box of toroids, I found a ferrite device that was about 1" (2.54cm) in outside diameter and wound as many turns of 14 AWG flexible wire onto it as would fit (about 6 bifilar turns) and measured it to have about 30 uH of inductance per winding.  This may not seem like much, but at 1 MHz, this represents about 180 ohms of reactance.   

In referring to Figure 5, above, you'll notice that it is pretty much identical to that of the output filter - except that there is only one section of filtering.  The capacitor values are different, too:  C1 and C2 are 0.1uF units that shunt residual RF getting through L1 to ground (the case) while C3 is a low-ESR electrolytic connected across the DC leads to help force any residual AC noise on the DC lead to common-mode.  Compared to the 180 ohms of reactance of the DC bifilar choke (at 1 MHz) a good-quality, monolithic ceramic capacitor like the 0.1uF units are likely to have well under an ohm of impedance and very little of the RF hash will remain after they do their job to bypass it to the chassis ground.

Figure 6: The DC input filter.  The capacitors (not visible) are mounted to the bottom side of the terminal strip, which serves as the RF "grounding" point to the case.  L1 is just visible. Click on the image for a larger version.

Because of the limited amount of room, only one inductor was used - although it would likely be possible to have crammed another in the limited space should the above filter have proved to be inadequate (it wasn't).

As can be seen in Figure 6, a small terminal strip is visible and to it is mounted C1-C3 (not visible as they are obscured by the strip itself).  The mounting point for this strip is the ground lug near the DC input cable and the center lug is the common point for C1 and C2.

An important point to mention is the fact that this inverter - like many - have their DC and AC lines isolated from the case - and that's also important here:  Because the DC has no connection to the inverter's metal case, ALL of the DC current passes through L1 of Figure 5 - but with both halves carrying the same current, the core is not magnetized:  Magnetizing the core would likely cause it to saturate and the result would be its effective inductance plummeting - possibly reducing its efficacy as an RF filter.  It is for this reason that a bifilar choke was used on the DC input as well.

As with the AC output, the "inverter" side of L1 of Figure 5 also lacks a common-mode capacitor, but this is well represented on the input of the inverter itself with its own, built-in capacitor.

Figure 7: The final arrangement of the added filtering components.  Liberal use of RTV (silicone adhesive) was used to stabilize the components as it works well, and can be removed should repairs/modifications be required.  On the left, a generous blob of RTV has been used to keep the terminal strip's lugs at the DC input from touching the inverter's bottom cover. Click on the image for a larger version.

Additional comments:

Figure 7 shows the final arrangement of the added components.  In the upper-left corner can be seen the components of the DC input filter with come clear RTV (silicone adhesive) added to the top of the terminal strip to insulate it and keep any metal parts of it from touching the bottom cover when it was reinstalled.

On the right side is the AC output filter and on the foreground can be seen L2, now with the "hot" terminals covered by heat-shrink tubing.  This choke was first attached "temporarily" to the inverter's end plate using instant (cyanoacrylate) glue - and then several large blobs of RTV were later added to permanently hold it in place.  Just above it can be seen the orange wires that connect L2 to L1 and these components were also stabilized with rather large blobs of RTV to keep them from "flapping in the breeze".  It's worth noticing that the original ferrite cylinder is still on the AC output connection (on the black and white wires) where it connects to L4 - mainly because there was still room for it, and its efficacy, such as it is, is likely only enhanced by the addition of the new filtering components. 

Did it work?

You might ask the question:  Did this filtering work?

The answer is yes.  Placing a portable shortwave radio next to either the DC or AC power leads from the inverter, one can't detect that it is running at all.  If the radio is placed right atop the inverter, some hash can be detected, but this is likely from direct radiation of magnetic fields from the inductors/transformers within, but detectable amounts do not appear to be emanating from DC and AC wires themselves - and that's the important part as they would otherwise be acting as antennas.

Perhaps the most important part of this modification is the fact that any bypass capacitors are placed on the "quiet" (not the inverter) side of the filtering inductances and that these bypass capacitors are connected, with short leads, to a large, common-point ground - namely the case of the inverter.  If any of the "ground" leads had been more than an inch or two long, it's likely that the impedance of it would have reduced the efficacy of the filtering - but the case, being a solid chunk of extruded aluminum, forms a nice, low-impedance tie point - effectively a single-point ground, preventing an RF current differential between the DC input and AC output leads.

* * *

Links to other articles about power supply noise reduction found at ka7oei.blogspot.com:

• A high-current DC noise filter for UPS or RV use.  What if you have an inverter or UPS that is large - as in the multi-kilowatt class?  This article describes quieting one such device.